Apparatus for transmitting and receiving a signal and method of transmitting and receiving a signal

ABSTRACT

According to one embodiment, a method of transmitting a broadcast signal includes: encoding physical layer pipe (PLP) data for delivering a service; mapping the encoded PLP data to symbols; building a signal frame based on the mapped symbols; modulating the signal frame according to an orthogonal frequency division multiplexing (OFDM) scheme; inserting a pilot symbol into a beginning part of the modulated signal frame; and transmitting the broadcast signal including the signal frame and the inserted pilot symbol. The pilot symbol comprises an effective portion, a cyclic prefix obtained by frequency-shifting a first portion of the effective portion, and a cyclic suffix obtained by frequency-shifting a second portion of the effective portion.

TECHNICAL FIELD

The present invention relates to a method for transmitting and receivinga signal and an apparatus for transmitting and receiving a signal, andmore particularly, to a method for transmitting and receiving a signaland an apparatus for transmitting and receiving a signal, which arecapable of improving data transmission efficiency.

BACKGROUND ART

As a digital broadcasting technology has been developed, users havereceived a high definition (HD) moving image. With continuousdevelopment of a compression algorithm and high performance of hardware,a better environment will be provided to the users in the future. Adigital television (DTV) system can receive a digital broadcastingsignal and provide a variety of supplementary services to users as wellas a video signal and an audio signal.

With the development of the digital broadcasting technology, arequirement for a service such as a video signal and an audio signal isincreased and the size of data desired by a user or the number ofbroadcasting channels is gradually increased.

DISCLOSURE OF INVENTION Technical Problem

As a digital broadcasting technology has been developed, users havereceived a high definition (HD) moving image. With continuousdevelopment of a compression algorithm and high performance of hardware,a better environment will be provided to the users in the future. Adigital television (DTV) system can receive a digital broadcastingsignal and provide a variety of supplementary services to users as wellas a video signal and an audio signal.

With the development of the digital broadcasting technology, arequirement for a service such as a video signal and an audio signal isincreased and the size of data desired by a user or the number ofbroadcasting channels is gradually increased.

Technical Solution

An object of the present invention is to provide a method fortransmitting and receiving a signal and an apparatus for transmittingand receiving a signal, which are capable of improving data transmissionefficiency.

Another object of the present invention is to provide a method fortransmitting and receiving a signal and an apparatus for transmittingand receiving a signal, which are capable of improving error correctioncapability of bits configuring a service.

Accordingly, the present invention is directed to a method fortransmitting and receiving a signal and an apparatus for transmittingand receiving a signal that substantially obviate one or more problemsdue to limitations and disadvantages of the related art.

To achieve these objects and other advantages and in accordance with thepurpose of the invention, as embodied and broadly described herein, amethod for transmitting a signal is provided. The method includesconverting (S110) a service stream to a physical layer pipe (PLP),allocating (S150) the PLP to a signal frame and arranging a preambleincluding a first pilot signal in a beginning part of the signal frame,converting (S160) the signal frame into a time domain according to anorthogonal frequency division multiplexing (OFDM) scheme, inserting(S170) a cyclic prefix obtained by frequency-shifting a first portion ofan useful portion of the first pilot signal and a cyclic suffix obtainedby frequency-shifting a second portion of said useful portion of thefirst pilot signal, into the first pilot signal and transmitting (S180)the signal frame including the first pilot signal over at least oneradio frequency (RF) channel.

The first pilot signal may have a structure according to the followingequation,

B=one part(A)·e ^(j2πf) ^(SH) ^(t)

C=another part(A)·e ^(j2πf) ^(SH) ^(t)

where, A denotes the valid portion of the first pilot signal, B denotesthe cyclic prefix, C denotes the cyclic suffix, and fSH denotes afrequency shift unit.

The first portion may be a foremost portion of the useful portion of thefirst pilot signal and the second portion may be a backmost portion ofthe useful portion of the first pilot signal.

In another aspect of the present invention, a method for receiving asignal is provided. The method includes receiving (S210) the signal in afirst frequency band, identifying (S220) a first pilot signal includinga cyclic prefix obtained by frequency-shifting a first portion of anuseful portion of the first pilot signal; and a cyclic suffix obtainedby frequency-shifting a second portion of the useful portion of thefirst pilot signal from the received signal, demodulating (S220) asignal frame including a physical layer pipe (PLP) to which a servicestream is converted, by an orthogonal frequency division multiplexing(OFDM) scheme, using information set in the first pilot signal, parsing(S230) the signal frame and obtaining the PLP, and obtaining (S240) theservice stream from the PLP.

The demodulating of the signal frame includes estimating a timing offsetand a fractional frequency offset of the received signal using thecyclic prefix and the cyclic suffix; and compensating for the estimatedoffsets.

In another aspect of the present invention, an apparatus fortransmitting a signal is provided. The apparatus includes a coding andmodulation unit (120) configured to error-correction-code a servicestream and to interleave the error-correction-coded service stream, aframe builder (130) configured to map bits of the interleaved servicestream to symbols of a physical layer pipe (PLP), allocate the PLP to asignal frame and arrange a preamble including a first pilot signal to abeginning part of the signal frame, a modulator (150 a) configured toconvert the signal frame into a time domain according to an orthogonalfrequency division multiplexing (OFDM) scheme and insert a cyclic prefixobtained by frequency-shifting a first portion of an useful portion ofthe first pilot signal and a cyclic suffix obtained byfrequency-shifting a second portion of the useful portion of the firstpilot signal, into the first pilot signal and a transmitter (160 a)configured to transmit the signal frame including the first pilot signalover at least one radio frequency (RF) channel.

In another aspect of the present invention, an apparatus for receiving asignal is provided. The apparatus includes first frequency band, ademodulator (220 a) configured to identify a first pilot signalincluding a cyclic prefix obtained by frequency-shifting a first portionof an useful portion of the first pilot signal and a cyclic suffixobtained by frequency-shifting a second portion of the useful portion ofthe first pilot signal from the received signal and demodulate a signalframe including a physical layer pipe (PLP), by an orthogonal frequencydivision multiplexing (OFDM) scheme, using information set in the firstpilot signal, a frame parser (240) configured to parse the signal frameand obtaining the PLP and demap symbols of the PLP to bits of a servicestream from the parsed signal frame and a decoding demodulator (250)configured to deinterleave the demapped bits of the service stream anddecode the deinterleaved bits of the service stream usingerror-correction-decoding scheme.

In another aspect of the present invention, a method is provided. Themethod includes allocating (S150) a service stream to a signal frame andarranging a preamble including a pilot signal in a beginning part of thesignal frame, modulating (S160) the signal frame, inserting (S170) acyclic prefix obtained by modifying a first portion of an useful portionof the pilot signal and a cyclic suffix obtained by modifying a secondportion of the useful portion of the pilot signal, into the pilot signaland transmitting (S180) the signal frame including the pilot signal.

In another aspect of the present invention, a method is provided. Themethod includes receiving (S210) the signal, identifying (S220) a signalframe from the received signal using a pilot signal including a cyclicprefix obtained by modifying a first portion of an useful portion of thepilot signal and a cyclic suffix obtained by modifying a second portionof the useful portion of first pilot signal and demodulating (S220) thesignal frame, parsing (S230) the signal frame and obtaining (S240) aservice stream from the parsed signal frame.

Advantageous Effects

According to the apparatus for transmitting and receiving the signal andthe method for transmitting and receiving the signal of the invention,if the data symbol configuring the PLP and the symbols configuring thepreamble are modulated in the same FFT mode, the probability that thedata symbol is detected by the preamble is low and the probability thatthe preamble is erroneously detected is reduced. If continuous wave (CW)interference is included like the analog TV signal, the probability thatthe preamble is erroneously detected by a noise DC component generatedat the time of correlation is reduced.

According to the apparatus for transmitting and receiving the signal andthe method for transmitting and receiving the signal of the invention,if the size of the FFT applied to the data symbol configuring the PLP islarger than that of the FFT applied to the preamble, the preambledetecting performance may be improved even in a delay spread channelhaving a length equal to or greater than that of the useful symbolportion A of the preamble. Since both the cyclic prefix (B) and thecyclic suffix (C) are used in the preamble, the fractional carrierfrequency offset can be estimated.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a view showing a signal frame for transmitting a service;

FIG. 2 is a view showing the structure of a first pilot signal P1 of thesignal frame;

FIG. 3 is a view showing a signaling window;

FIG. 4 is a schematic view showing an embodiment of an apparatus fortransmitting a signal;

FIG. 5 is a view showing an example of an input processor 110;

FIG. 6 is a view showing an embodiment of a coding and modulation unit;

FIG. 7 is a view showing an embodiment of a frame builder;

FIG. 8 is a view showing a first example of a ratio of symbols whenmappers 131 a and 131 b perform hybrid symbol mapping;

FIG. 9 is a view showing a second example of a ratio of symbols when themappers 131 a and 131 b perform hybrid symbol mapping;

FIG. 10 is a view showing the number of symbols and bit number per cellword according to a symbol mapping scheme in an LDPC normal mode;

FIG. 11 is a view showing another example of the number of symbolsaccording to a symbol mapping scheme in an LDPC normal mode;

FIG. 12 is a view showing another example of the number of symbolsaccording to a symbol mapping scheme in an LDPC normal mode;

FIG. 13 is a view showing the number of symbols according to a symbolmapping scheme in an LDPC short mode;

FIG. 14 is a view showing an example of the number of symbols accordingto a symbol mapping scheme in an LDPC short mode;

FIG. 15 is a view showing another example of the number of symbolsaccording to a symbol mapping scheme in an LDPC short mode;

FIG. 16 is a view showing an embodiment of each of the symbol mappers131 a and 131 b shown in FIG. 7;

FIG. 17 is a view showing another embodiment of each of the symbolmappers 131 a and 131 b;

FIG. 18 is a view showing another embodiment of the symbol mapper;

FIG. 19 is a view showing another embodiment of each of the symbolmappers 131 a and 131 b;

FIG. 20 is a view showing the concept of interleaving of bits by bitinterleavers 1312 a and 1312 b;

FIG. 21 is a view showing a first example of the number of rows andcolumns of memories of the bit interleavers 1312 a and 1312 b accordingto the types of symbol mappers 1315 a and 1315 b;

FIG. 22 is a view showing a second example of the number of rows andcolumns of the memories of the bit interleavers 1312 a and 1312 baccording to the types of the symbol mappers 1315 a and 1315 b;

FIG. 23 is a diagram showing the concept of another embodiment ofinterleaving of a bit interleaver;

FIG. 24 is a view showing another embodiment of bit interleaving;

FIG. 25 is a view showing another embodiment of bit interleaving;

FIG. 26 is a view showing the concept of demultiplexing of input bits ofdemurs 1313 a and 1313 b;

FIG. 27 is a view showing an embodiment of demultiplexing an inputstream by the demux;

FIG. 28 is a view showing an example of a demultiplexing type accordingto a symbol mapping method;

FIG. 29 is a view showing an embodiment of demultiplexing an input bitstream according to a demultiplexing type;

FIG. 30 is a view showing a demultiplexing type which is determinedaccording to a code rate of an error correction coding and a symbolmapping method;

FIG. 31 is a view showing an example of expressing the demultiplexingmethod by an equation;

FIG. 32 is a view showing an example of mapping a symbol by a symbolmapper;

FIG. 33 is a view showing an example of a multi-path signal coder;

FIG. 34 is a view showing an embodiment of a modulator;

FIG. 35 is a view showing an embodiment of an analog processor 160;

FIG. 36 is a view showing an embodiment of a signal receiving apparatuscapable of receiving a signal frame;

FIG. 37 is a view showing an embodiment of a signal receiver;

FIG. 38 is a view showing an embodiment of a demodulator;

FIG. 39 is a view showing a multi-path signal decoder;

FIG. 40 is a view showing an embodiment of a frame parser;

FIG. 41 is a view showing an embodiment of each of symbol demappers 247a and 247 p;

FIG. 42 is a view showing another embodiment of each of the symboldemappers 247 a and 247 p;

FIG. 43 is a view showing another embodiment of each of the symboldemappers 247 a and 247 p;

FIG. 44 is a view showing another embodiment of each of the symboldemappers 247 a and 247 p;

FIG. 45 is a view showing an embodiment of multiplexing a demultiplexedsub stream;

FIG. 46 is a view showing an example of a decoding and demodulationunit;

FIG. 47 is a view showing an embodiment of an output processor;

FIG. 48 is a view showing another embodiment of a signal transmittingapparatus for transmitting a signal frame;

FIG. 49 is a view showing another embodiment of a signal receivingapparatus for receiving a signal frame;

FIG. 50 is a view showing an embodiment of the structure of a firstpilot signal;

FIG. 51 is a view showing an embodiment of detecting a preamble signalshown in FIG. 50 and estimating a timing offset and a frequency offset;

FIG. 52 is a view showing another embodiment of the structure of thefirst pilot signal;

FIG. 53 is a view showing an embodiment of detecting the first pilotsignal shown in FIG. 52 and measuring a timing offset and a frequencyoffset;

FIG. 54 is a view showing an embodiment of detecting the first pilotsignal and measuring a timing offset and a frequency offset using thedetected result;

FIG. 55 is a view showing an embodiment of a method of transmitting asignal;

FIG. 56 is a view showing an embodiment of a method of receiving asignal; and

FIG. 57 is a flowchart illustrating an embodiment of identifying a firstpilot signal and estimating an offset in a demodulating process.

BEST MODE FOR CARRYING OUT THE INVENTION

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. Wherever possible, the same reference numbers will be usedthroughout the drawings to refer to the same or like parts.

In the following description, the term “service” is indicative of eitherbroadcast contents which can be transmitted/received by the signaltransmission/reception apparatus, or content provision.

Prior to the description of an apparatus for transmitting and receivinga signal according to an embodiment of the present invention, a signalframe which is transmitted and received by the apparatus fortransmitting and receiving the signal according to an embodiment of thepresent invention will be described.

FIG. 1 shows a signal frame for transmitting a service according to anembodiment of the present invention.

The signal frame shown in FIG. 1 shows an exemplary signal frame fortransmitting a broadcast service including audio/video (A/V) streams. Inthis case, a single service is multiplexed in time- andfrequency-channels, and the multiplexed service is transmitted. Theabove-mentioned signal transmission scheme is called a time-frequencyslicing (TFS) scheme. Compared with the case in which a single serviceis transmitted to only one radio frequency (RF) band, the signaltransmission apparatus according to an embodiment of he presentinvention transmits the signal service via at least one RF band(possibly several RF bands), such that it can acquire a statisticalmultiplexing gain capable of transmitting many more services. The signaltransmission/reception apparatus transmits/receives a single serviceover several RF channels, such that it can acquire a frequency diversitygain.

First to third services (Services 1˜3) are transmitted to four RF bands(RF1˜RF4). However, this number of RF bands and this number of serviceshave been disclosed for only illustrative purposes, such that othernumbers may also be used as necessary. Two reference signals (i.e., afirst pilot signal (P1) and a second pilot signal (P2)) are located atthe beginning part of the signal frame. For example, in the case of theRF1 band, the first pilot signal (P1) and the second pilot signal (P2)are located at the beginning part of the signal frame. The RF1 bandincludes three slots associated with the Service 1, two slots associatedwith the Service 2, and a single slot associated with the Service 3.Slots associated with other services may also be located in other slots(Slots 4-17) located after the single slot associated with the Service3.

The RF2 band includes a first pilot signal (P1), a second pilot signal(P2), and other slots 13˜17. In addition, the RF2 band includes threeslots associated with the Service 1, two slots associated with theService 2, and a single slot associated with the Service 3.

The Services 1˜3 are multiplexed, and are then transmitted to the RF3and RF4 bands according to the time-frequency slicing (TFS) scheme. Themodulation scheme for signal transmission may be based on an orthogonalfrequency division multiplexing (OFDM) scheme.

In the signal frame, individual services are shifted to the RF bands (inthe case that there is a plurality of the RF bands in the signal frame)and a time axis.

If signal frames equal to the above signal frame are successivelyarranged in time, a super-frame can be composed of several signalframes. A future extension frame may also be located among the severalsignal frames. If the future extension frame is located among theseveral signal frames, the super-frame may be terminated at the futureextension frame.

FIG. 2 shows a first pilot signal (P1) contained in the signal frame ofFIG. 1 according to an embodiment of the present invention.

The first pilot signal P1 and the second pilot signal P2 are located atthe beginning part of the signal frame. The first pilot signal P1 ismodulated by a 2K FFT mode, and may be transmitted simultaneously whileincluding a ¼ guard interval. In FIG. 2, a band of 7.61 Mhz of the firstpilot signal P1 includes a band of 6.82992 Mhz. The first pilot signaluses 256 carriers from among 1705 active carriers. A single activecarrier is used for every 6 carriers on average. Data-carrier intervalsmay be irregularly arranged in the order of 3, 6, and 9. In FIG. 2, asolid line indicates the location of a used carrier, a thin dotted lineindicates the location of an unused carrier, and a chain line indicatesa center location of the unused carrier. In the first pilot signal, theused carrier can be symbol-mapped by a binary phase shift keying (BPSK),and a pseudo-random bit sequence (PRBS) can be modulated. The size of aFFT used for the second pilot signal can be indicated by several PRBSs.

The signal reception apparatus detects a structure of a pilot signal,and recognizes a time-frequency slicing (TFS) using the detectedstructure. The signal reception apparatus acquires the FFT size of thesecond pilot signal, compensates for a coarse frequency offset of areception signal, and acquires time synchronization.

In the first pilot signal, a signal transmission type and a transmissionparameter may be set.

The second pilot signal P2 may be transmitted with a FFT size and aguard interval equal to those of the data symbol. In the second pilotsignal, a single carrier is used as a pilot carrier at intervals ofthree carriers. The signal reception apparatus compensates for a finefrequency synchronization offset using the second pilot signal, andperforms fine time synchronization. The second pilot signal transmitsinformation of a first layer (L1) from among Open SystemsInterconnection (OSI) layers. For example, the second pilot signal mayinclude a physical parameter and frame construction information. Thesecond pilot signal transmits a parameter value by which a receiver canaccess a Physical Layer Pipe (PLP) service stream.

L1 (Layer 1) information contained in the second pilot signal P2 is asfollows.

The Layer-1 (L1) information includes a length indicator indicating thelength of data including the L1 information, such that it can easily usethe signaling channels of Layers 1 and 2 (L1 and L2). The Layer-1 (L1)information includes a frequency indicator, a guard-interval length, amaximum number of FEE (Forward Error Correction) blocks for each framein association with individual physical channels, and the number ofactual FEC blocks to be contained in the FEC block buffer associatedwith a current/previous frame in each physical channel. In this case,the frequency indicator indicates frequency information corresponding tothe RF channel.

The Layer-1 (L1) information may include a variety of information inassociation with individual slots. For example, the Layer-1 (L1)information includes the number of frames associated with a service, astart address of a slot having the accuracy of an OFDM carrier containedin an OFDM symbol, a length of the slot, slots corresponding to the OFDMcarrier, the number of bits padded in the last OFDM carrier, servicemodulation information, service mode rate information, andMulti-Input-Multi-Output (MIMO) scheme information.

The Layer-1 (L1) information may include a cell ID, a flag for servicelike notification message service (e.g., an emergency message), thenumber of current frames, and the number of additional bits for futureuse. In this case, the cell ID indicates a broadcast area transmitted bya broadcast transmitter

The second pilot signal P2 is adapted to perform channel estimation fordecoding a symbol contained in the P2 signal. The second pilot signal P2can be used as an initial value for channel estimation for the next datasymbol. The second pilot signal P2 may also transmit Layer-2 (L2)information. For example, the second pilot signal is able to describeinformation associated with the transmission service in Layer-2 (L2)information. The signal transmission apparatus decodes the second pilotsignal, such that it can acquire service information contained in thetime-frequency slicing (TFS) frame and can effectively perform thechannel scanning. Meanwhile, this Layer-2 (L2) information may beincluded in a specific PLP of the TFS frame. According to anotherinstance, L2 information can be included in a specific PLP, and theservice description information also can be transmitted in the specificPLP.

For example, the second pilot signal may include two OFDM symbols of the8k FFT mode. Generally, the second pilot signal may be any one of asingle OFDM symbol of the 32K FFT mode, a single OFDM symbol of the 16kFFT mode, two OFDM symbols of the 8k FFT mode, four OFDM symbols of the4k FFT mode, and eight OFDM symbols of the 2k FFT mode.

In other words, a single OFDM symbol having the size of a large FFT orseveral OFDM symbols, each of which has the size of a small FFT, may becontained in the second pilot signal P2, such that capacity capable ofbeing transmitted to the pilot can be maintained.

If information to be transmitted to the second pilot signal exceedscapacity of the OFDM symbol of the second pilot signal, OFDM symbolsafter the second pilot signal can be further used. L1 (Layer1) and L2(Layer2) information contained in the second pilot signal iserror-correction-coded and is then interleaved, such that data recoveryis carried out although an impulse noise occurs.

As described the above, L2 information can also be included in aspecific PLP conveying the service description information.

FIG. 3 shows a signaling window according to an embodiment of thepresent invention. The time-frequency slicing (TFS) frame shows anoffset concept of the signaling information. Layer-1 (L1) informationcontained in the second pilot signal includes frame constructioninformation and physical layer information required by the signalreception apparatus decoding the data symbol. Therefore, if informationof the following data symbols located after the second pilot signal, iscontained in the second pilot signal, and the resultant second pilotsignal is transmitted, the signal reception apparatus may be unable toimmediately decode the above following data symbols due to a decodingtime of the second pilot signal.

Therefore, as shown in FIG. 3, the L1 information contained in thesecond pilot signal (P2) includes information of a single time-frequencyslicing (TFS) frame size, and includes information contained in thesignaling window at a location spaced apart from the second pilot signalby the signaling window offset.

In the meantime, in order to perform channel estimation of a data symbolconstructing the service, the data symbol may include a scatter pilotand a continual pilot.

The signal transmission/reception system capable oftransmitting/receiving signal frames shown in FIGS. 1˜3 will hereinafterbe described. Individual services can be transmitted and received overseveral RF channels. A path for transmitting each of the services or astream transmitted via this path is called a PLP. The PLP may bedistributed among the timely-divided slots in several RF channels or asingle RF band.

This signal frame can convey the timely-divided PLP in at least one RFchannel. In other word, a single PLP can be transferred through at leastone RF channel with timely-divided regions. Hereinafter the signaltransmission/reception systems transmitting/receiving a signal frame viaat least one RF band will be disclosed.

FIG. 4 is a block diagram illustrating an apparatus for transmitting asignal according to one embodiment of the present invention. Referringto FIG. 4, the signal transmission apparatus includes an input processor110, a coding and modulation unit 120, a frame builder 130, a MIMO/MISOencoder 140, a plurality of modulators (150 a, . . . , 150 r) of theMIMO/MISO encoder 140, and a plurality of analog processors (160 a, . .. , 160 r).

The input processor 110 receives streams equipped with several services,generates P number of baseband frames (P is a natural number) whichincludes modulation- and coding-information corresponding totransmission paths of the individual services, and outputs the P numberof baseband frames.

The coding and modulation unit 120 receives baseband frames from theinput processor 110, performs the channel coding and interleaving oneach of the baseband frames, and outputs the channel coding andinterleaving result.

The frame builder 130 forms frames which transmit baseband framescontained in P number of PLPs to R number of RF channels (where R is anatural number), splits the formed frames, and outputs the split framesto paths corresponding to the R number of RF channels. Several servicesmay be multiplexed in a single RF channel in time. The signal framesgenerated from the frame builder 140 may include a time-frequencyslicing (TFS) structure in which the service is multiplexed in time- andfrequency-domains.

The MIMO/MISO encoder 140 encodes signals to be transmitted to the Rnumber of RF channels, and outputs the coded signals to pathscorresponding to A number of antennas (where A is a natural number). TheMIMO/MISO encoder 140 outputs the coded signal in which a single to betransmitted to a single RF channel is encoded to the A number ofantennas, such that a signal is transmitted/received to/from a MEMO(Multi-Input-Multi-Output) or MISO (Multi-Input-Single-Output)structure.

The modulators (150 a, . . . , 150 r) modulate frequency-domain signalsentered via the path corresponding to each RF channel into time-domainsignals. The modulators (150 a, . . . , 150 r) modulate the inputsignals according to an orthogonal frequency division multiplexing(OFDM) scheme, and outputs the modulated signals.

The analog processors (160 a, . . . , 160 r) convert the input signalsinto RF signals, such that the RF signals can be outputted to the RFchannels.

The signal transmission apparatus according to this embodiment mayinclude a predetermined number of modulators (150 a, . . . 150 r)corresponding to the number of RF channels and a predetermined number ofanalog processors (160 a, . . . , 160 r) corresponding to the number ofRF channels. However, in the case of using the MIMO scheme, the numberof analog processors must be equal to the product of R (i.e., the numberof RF channels) and A (i.e., the number of antennas).

FIG. 5 is a block diagram illustrating an input processor 110 accordingto an embodiment of the present invention. Referring to FIG. 5, theinput processor 110 includes the first stream multiplexer 111 a, thefirst service splitter 113 a, and a plurality of first baseband (BB)frame builders (115 a, . . . 115 m). The input processor 110 includes asecond stream multiplexer 111 b, a second service splitter 113 b, and aplurality of second baseband (BB) frame builders (115 n, . . . , 115 p).

For example, the first stream multiplexer 111 a receives several MPEG-2transport streams (TSs), multiplexes the received MPEG-2 TS streams, andoutputs the multiplexed MPEG-2 TS streams. The first service splitter113 a receives the multiplexed streams, splits the input streams ofindividual services, and outputs the split streams. As described above,provided that the service transmitted via a physical-channel path iscalled a PLP, the first service splitter 113 a splits the service to betransmitted to each PLP, and outputs the split service.

The first BB frame builders (115 a, . . . , 115 m) build data containedin a service to be transmitted to each PLP in the form of a specificframe, and output the specific-frame-formatted data. The first BB framebuilders (115 a, 115 m) build a frame including a header and payloadequipped with service data. The header of each frame may include modeinformation based on the modulation and encoding of the service data,and a counter value based on a clock rate of the modulator tosynchronize input streams.

The second stream multiplexer 111 b receives several streams,multiplexes input streams, and outputs the multiplexed streams. Forexample, the second stream multiplexer 111 b may multiplex InternetProtocol (IP) streams instead of the MPEG-2 TS streams. These streamsmay be encapsulated by a generic stream encapsulation (GSE) scheme. Thestreams multiplexed by the second stream multiplexer 111 b may be anyone of streams. Therefore, the above-mentioned streams different fromthe MPEG-2 TS streams are called generic streams (GS streams).

The second service splitter 113 b receives the multiplexed genericstreams, splits the received generic streams according to individualservices (i.e., PLP types), and outputs the split GS streams.

The second BB frame builders (115 n, . . . , 115 p) build service datato be transmitted to individual PLPs in the form of a specific frameused as a signal processing unit, and output the resultant service data.The frame format built by the second BB frame builders (115 n, . . . ,115 p) may be equal to that of the first BB frame builders (115 a, . . ., 115 m) as necessary. If required, another embodiment may also beproposed. In another embodiment, the frame format built by the second BBframe builders (115 n, 115 p) may be different from that of the first BBframe builders (115 a, . . . , 115 m). The MPEG2 TS header furtherincludes a Packet Syncword which is not contained in the GS stream,resulting in the occurrence of different headers.

FIG. 6 is a block diagram illustrating a coding and modulation unitaccording to an embodiment of the present invention. The coding andmodulation unit includes a first interleaver 123, a second encoder 125,and a second interleaver 127.

The first encoder 121 acts as an outer coder of the input basebandframe, and is able to perform the error correction encoding. The firstencoder 121 performs the error correction encoding of the input basebandframe using a Bose-Chaudhuri-Hocquenghem (BCH) scheme. The firstinterleaver 123 performs interleaving of the encoded data, such that itprevents a burst error from being generated in a transmission signal.The first interleaver 123 may not be contained in the above-mentionedembodiment.

The second encoder 125 acts as an inner coder of either the output dataof the first encoder 121 or the output data of the first interleaver123, and is able to perform the error correction encoding. A low densityparity bit (LDPC) scheme may be used as an error correction encodingscheme. The second interleaver 127 mixes the error-correction-encodeddata generated from the second encoder 125, and outputs the mixed data.The first interleaver 123 and the second interleaver 127 are able toperform interleaving of data in units of a bit.

The coding and modulation unit 120 relates to a single PLP stream. ThePLP stream is error-correction-encoded and modulated by the coding andmodulation unit 120, and is then transmitted to the frame builder 130.

FIG. 7 is a block diagram illustrating a frame builder according to anembodiment of the present invention. Referring to FIG. 7, the framebuilder 130 receives streams of several paths from the coding andmodulation unit 120, and arranges the received streams in a singlesignal frame. For example, the frame builder may include a first mapper131 a and a first time interleaver 132 a in a first path, and mayinclude a second mapper 131 b and a second time interleaver 132 b in asecond path. The number of input paths is equal to the number of PLPsfor service transmission or the number of streams transmitted via eachPLP.

The first mapper 131 a performs mapping of data contained in the inputstream according to the first symbol mapping scheme. For example, thefirst mapper 131 a may perform mapping of the input data using a QAMscheme (e.g., 16 QAM, 64 QAM, and 256 QAM).

If the first mapper 131 a performs mapping of the symbol, the input datamay be mapped to several kinds of symbols according to several symbolmapping schemes. For example, the first mapper 131 a classifies theinput data into a baseband-frame unit and a baseband-frame sub-unit.Individual classified data may be hybrid-symbol-mapped by at least twoQAM schemes (e.g., 16 QAM and 64 QAM). Therefore, data contained in asingle service may be mapped to symbols based on different symbolmapping schemes in individual intervals.

The first time interleaver 132 a receives a symbol sequence mapped bythe first mapper 131 a, and is able to perform the interleaving in atime domain. The first mapper 131 a maps data, which is contained in theerror-corrected frame unit received from the coding and modulation unit120, into symbols. The first time interleaver 132 a receives the symbolsequence mapped by the first mapper 131 a, and interleaves the receivedsymbol sequence in units of the error-corrected frame.

In this way, the p-th mapper 131 p or the p-th time interleaver 132 preceives service data to be transmitted to the p-th PLP, maps theservice data into symbols according to the p-th symbol mapping scheme.The mapped symbols can be interleaved in a time domain. It should benoted that this symbol mapping scheme and this interleaving scheme areequal to those of the first time interleaver 132 a and the first mapper131 a.

The symbol mapping scheme of the first mapper 131 a may be equal to ordifferent from that of the p-th mapper 131 p. The first mapper 131 a andthe p-th mapper 131 p are able to map input data to individual symbolsusing the same or different hybrid symbol mapping schemes.

Data of the time interleavers located at individual paths (i.e., servicedata interleaved by the first time interleaver 132 a and service data tobe transmitted to R number of RE channels by the p-th time interleaver132 p) is interleaved, such that the physical channel allows the abovedata to be interleaved over several RF channels.

In association with streams received in as many paths as the number ofPLPs, the TFS frame builder 133 builds the TFS signal frame such as theabove-mentioned signal frame, such that the service is time-shiftedaccording to RF channels. The TFS frame builder 133 splits service datareceived in any one of paths, and outputs the service data split intodata of the R number of RF bands according to a signal schedulingscheme.

The TFS frame builder 133 receives the first pilot signal and the secondpilot signal from the signaling information unit (denoted by Ref/PLsignal) 135, arranges the first and second pilot signals in the signalframe, and inserts the signaling signal (L1 and L2) of theabove-mentioned physical layer in the second pilot signal. In this case,the first and second pilot signals are used as the beginning signals ofthe signal frame contained in each RF channel from among the TFS signalframe received from the signaling information unit (Ref/PL signal) 135.As shown in FIG. 2, the first pilot signal may include a transmissiontype and basic transmission parameters, and the second pilot signal mayinclude a physical parameter and frame construction information. Also,the second pilot signal includes a L1 (Layer 1) signaling signal and aL2 (Layer 2) signaling signal.

The R number of frequency interleavers (137 a, . . . , 137 r) interleaveservice data, to be transmitted to corresponding RF channels of the TFSsignal frame, in a frequency domain. The frequency interleavers (137 a,. . . , 137 r) can interleave the service data at a level of data cellscontained in an OFDM symbol.

Therefore, service data to be transmitted to each RF channel in the TFSsignal frame is frequency-selective-fading-processed, such that it maynot be lost in a specific frequency domain.

FIG. 8 is a view showing a first example of a ratio of symbols when themappers 131 a and 131 b perform hybrid symbol mapping. This Figure showsthe number of bits transmitted by one sub carrier (cell) if errorcorrection coding is performed by the coding and modulation unit in anormal mode (the length of the error-correction-coded code is 64800bits) of LDPC error correction coding mode.

For example, if the mappers 131 a and 131 b perform symbol mapping using256QAM, 64800 bits are mapped to 8100 symbols. If the mappers 131 a and131 b perform hybrid symbol mapping (Hyb 128-QAM) using 256QAM and 64QAMwith a ratio of 3:2, the number of symbols mapped by 256QAM is 4860 andthe number of symbols mapped by 64QAM is 4320. The number of transmittedbits per sub carrier (cell) is 7.0588.

If a symbol mapping method of 64QAM is used, input data may be mapped to10800 symbols and six bits per cell may be transmitted. If data ismapped to the symbols by a hybrid symbol mapping method of 64QAM and16QAM (64QAM:16QAM=3:2, Hyb32-QAM), five bits may be transmitted by onesub carrier (cell).

If data is mapped to symbols by the 16QAM method, the data is mapped to16200 symbols, each of which is used to transmit four bits.

Similarly, if data is mapped to symbols by a hybrid symbol mappingmethod of 16QAM and QPSK (16QAM:QPSK=2:3, Hyb8-QAM), three bits may betransmitted by one sub carrier (cell).

If data is mapped to symbols by a QPSK method, the data may be mapped to32400 symbols, each of which is used to transmit two bits.

FIG. 9 shows symbol mapping methods of error-corrected data by LDPCerror correction coding method of a short mode (the length of theerror-correction-coded code is 16200 bits), which are equal to thesymbol mapping methods of FIG. 8, and the numbers of bits per subcarrier according to the symbol mapping methods.

The numbers of bits transmitted by the sub carrier is equal to those ofthe normal mode (64800 bits) according to the symbol mapping methodssuch as 256QAM, Hyb 128QAM, 64-QAM, Hyb 32-QAM, 16QAM, Hyb8-QAM andQPSK, but the total numbers of symbols transmitted are different fromthose of the normal mode. For example, 16200 bits are transmitted by2025 symbols in 256QAM, 16200 bits are transmitted by 1215 symbolsaccording to 256QAM and 1080 symbols according to 64QAM (total 2295symbols) in Hyb 128-QAM.

Accordingly, a data transmission rate per sub carrier (cell) for eachPLP may be adjusted according to a hybrid symbol mapping method or asingle symbol mapping method.

FIG. 10 is a view showing the number of symbols and bit number per cellword according to a symbol mapping method in an LDPC normal mode. If aTFS signal frame includes at least one RF channel, symbols configuring aspecific PLP can be uniformly allocated to RF channels. The locations ofthe PLP symbols allocated to the RF channels can be more efficientlyaddressed. Accordingly, when the signal receiving apparatus selects theRF channels, the bits used for addressing the specific PLP can bereduced.

In this drawing, a symbol mapping method represented by 256-QAMindicates a method of mapping bits configuring a singleerror-correction-coded block to symbols with a ratio of256QAM:64QAM=8:1. According to this symbol mapping method, the number ofthe bits in a single error-correction-coded block by the 256-QAM methodis 57600, the number of the bits in a single error-correction-codedblock by the 256-QAM method is 1200, the number of total symbols in theblock is 8400, and the bit number per cell word is 7.714285714.

A symbol mapping method represented by Hyb 128-QAM indicates a method ofmapping bits configuring a single error-correction-coded block tosymbols with a ratio of 256QAM:64QAM=8:7. According to the Hyb 128-QAMsymbol mapping method, the number of total symbols in a singleerror-correction-encoding block is 9600, and the bit number per cellword is 6.75.

According to a symbol mapping method represented by 64QAM, the number oftotal symbols in a single error-correction-encoding block is 10800 andthe bit number per cell word is 6.

A symbol mapping method represented by Hyb 32-QAM indicates a method ofmapping bits configuring a single error-correction-coded block tosymbols with a ratio of 64QAM:32QAM=5:4. According to the Hyb 32-QAMsymbol mapping method, the number of total symbols in theerror-correction-coded block is 13200, and the bit number per cell wordis 4.9090909.

A symbol mapping method represented by 16 QAM indicates a method ofmapping bits configuring a single error-correction-coded block tosymbols with a ratio of 16QAM:QPSK=1:8. According to the 16 QAM symbolmapping method, the number of total symbols in oneerror-correction-coded block is 15600. and the bit number per cell wordis 4.153846154.

A symbol mapping method represented by Hyb S-QAM indicates a method ofmapping bits configuring a single error-correction-coded block tosymbols with a ratio of 16QAM:QPSK=2:1. According to the Hyb 8-QAMsymbol mapping method, the number of total symbols in oneerror-correction-coded block is 21600, and the bit number per cell wordis 3.

According to a symbol mapping method represented by QPSK, the number oftotal symbols in one error-correction-coded block is 32400 and the bitnumber per cell word is 2.

When the symbols configuring the PLP are allocated to the RF channels,the diversity gain of the frequency domain can be maximized when thenumbers of the symbols allocated to the respective RF channels areequal. If a maximum of six RF channels is considered, the lowest commonmultiple of 1 to 6 is 60 and the greatest common divisor of the numbersof symbols mapped to one error correction coded-block is 1200.Accordingly, if the integral multiple of 1200/60=20 symbols is allocatedto each of the RF channels, the symbols can be uniformly allocated toall the RF channels. At this time, if 20 symbols are considered as onegroup and the group is addressed, the addressing overhead of log2(20)4.32 bits can be reduced compared with the case the symbols areaddressed one by one.

FIG. 11 is a view showing another example of the number of symbolsaccording to a symbol mapping method in an LDPC normal mode. In theexample of this drawing, a 256-QAM method using 256QAM and 64QAM symbols(256QAM:64 QAM=4:1), a Hyb 128-QAM method using 256QAM and 64QAM symbol(256QAM:64QAM=8:7), a 64QAM method, a Hyb 32-QAM method using 64QAM and8QAM symbols (54QAM:8QAM=3:2), a 16 QAM method using 16QAM and QPSKsymbols (16QAM:QPSK=1:14), a Hyb 8-QAM method using 16QAM:QPSK=2:1 and aQPSK method were used as the symbol mapping method. The greatest commondivisor (GCD) of the numbers of total symbols of an error correctioncoded block (normal mode) according to the symbol mapping methods is720. Accordingly, if the integral multiple of 12 (=720/60) symbols isallocated to each of the RF channels, the symbols can be uniformlyallocated to all the RF channels. At this time, if 12 symbols areconsidered as one group and the group is addressed, the addressingoverhead of log 2(12)3.58 bits can be reduced compared with the case thesymbols are addressed one by one. The signal receiving apparatus cancollect the allocated PLP symbols by the addressing scheme and obtain aPLP service stream.

FIG. 12 is a view showing another example of the number of symbolsaccording to a symbol mapping method in an LDPC normal mode. In theexample of this drawing, a 256-QAM scheme, a Hyb 128-QAM scheme, a 64QAMscheme, a Hyb 32-QAM scheme, a 16 QAM scheme, a Hyb 8-QAM scheme and aQPSK scheme were used as the symbol mapping method. The 256QAM symbolmapping method uses 256QAM and 64QAM symbols (256QAM:64 QAM=44:1) andthe Hyb 128-QAM symbol mapping method uses 256QAM and 64QAM symbols(256QAM:64QAM=28:17). The Hyb 32-QAM method uses 64QAM and 8QAM symbols(64QAM:8QAM=3:2), the 16QAM symbol mapping method uses 16QAM and QPSKsymbols (16QAM: QPSK=1:14), and the Hyb 8-QAM symbol mapping method uses16QAM and QPSK symbols (16QAM:QPSK=2:1). The GCD of the numbers of totalsymbols of an error correction coded block (normal mode) according tothe symbol mapping methods is 240. Accordingly, if the integral multipleof 240/60=4 symbols is allocated to each of the RF channels, the symbolscan be uniformly allocated to all the RF channels. At this time, if foursymbols are considered as one group and the group is addressed, theaddressing overhead of log 2(4)2 bits can be reduced compared with thecase where the symbols are addressed one by one. Accordingly, even whenthe number of RE channels is any one of 1 to 6 in the signal frame, thePLP symbols can be uniformly allocated to the RF channels.

FIG. 13 is a view showing the number of symbols according to a symbolmapping method in an LDPC short mode. As described above, if symbolmapping is performed according to this example, the PLP symbols can beuniformly allocated to the RF channels and the overhead of the PLPsymbol addressing can be reduced. The symbol mapping methods shown inthis drawing are equal to those shown in FIG. 10. However, since the bitnumber of the LDPC short mode is different from that of the normal mode,the GCD of the numbers of total symbols of an error correction codedblock (short mode) according to the symbol mapping methods is 300,unlike to FIG. 10. Accordingly, if the integral multiple of 300/60=5symbols is allocated to each of the RF channels, the symbols can beuniformly allocated to all the RF channels. At this time, if fivesymbols are considered as one group and the group is addressed, theaddressing overhead of log 2(5) bits can be reduced compared with thecase where the symbols are addressed one by one. Accordingly, in thisembodiment, the addressing bits are saved by log 2(5) bits when thedivided PLP symbols are addressed.

FIG. 14 is a view showing an example of the number of symbols accordingto a symbol mapping method in an LDPC short mode. The symbol mappingmethods of this drawing are equal to those shown in FIG. 11. In thisexample, the GCD of the numbers of total symbols of an error correctioncoded block (short mode) according to the symbol mapping methods is 180,which may be used for PLP symbol allocation of one RF channel and theaddressing of the allocated symbols. In this embodiment, the addressingbits are saved by log 2(3) bits.

FIG. 15 is a view showing another example of the number of symbolsaccording to a symbol mapping method in an LDPC short mode. The symbolmapping methods of this drawing are equal to those shown in FIG. 12. Inthis example, the GCD of the numbers of total symbols of an errorcorrection coded block (short mode) according to the symbol mappingmethods is 60. In this embodiment, the addressing bits are saved by log2(1) bits (that is, the addressing bit is not saved).

FIG. 16 is a view showing an example of each of the symbol mappers 131 aand 131 b shown in FIG. 7. Each of the symbol mappers 131 a and 131 bincludes a first order mapper 1315 a, a second order mapper 131 b, asymbol merger 1317 and an error correction block merger 1318.

The bit stream parser 1311 receives the PLP service stream from thecoding and modulation unit and splits the received service stream.

The first order symbol mapper 1315 a maps the bits of the service streamsplit by a higher order symbol mapping method to symbols. The secondorder symbol mapper 1315 b maps the bits of the service stream split bya lower order symbol mapping method to symbols. For example, in theabove example, the first order symbol mapper 1315 a may map the bitstream to symbols according to 256QAM and the second order symbol mapper1315 b may map the bit stream to symbols according to 64QAM.

The symbol merger 1317 merges the symbols output from the symbol mappers1315 a and 1315 b to one symbol stream and outputs the symbol stream.The symbol merger 1317 may output the symbol stream included in one PLP.

The error correction block merger 1318 may output one symbol streammerged by the symbol merger 1317 in the error-correction-coded codeblock unit. The error correction block merger 1318 may output a symbolblock such that the error-correction-coded code blocks are uniformlyallocated to at least one RF band of the TFS signal frame. The errorcorrection block merger 1318 may output the symbol block such that thelength of the symbol block of the error-correction-coded block of anormal mode is equal to that of the symbol block of theerror-correction-coded block of a short mode. For example, four symbolblocks of the error-correction-coded block of the short mode may bemerged to one symbol block.

The error correction block merger 1318 may split the symbol streamaccording to a common multiple of the number of RF bands such thatsignal frame builder uniformly arranges the symbols to the RF bands. Ifthe maximum number of RF bands in the signal frame is 6, the errorcorrection block merger 1318 outputs the symbol block such that thetotal number of symbols can be divided by 60 which is a common multipleof 1, 2, 3, 4, 5 and 6.

The symbols included in the output symbol block may be arranged to beuniformly allocated to the six RF bands. Accordingly, although an errorcorrection mode according to a code rate and a symbol mapping method arecombined, the symbols configuring the PLP are uniformly allocated to theRF bands.

FIG. 17 is a view showing another embodiment of each of the symbolmappers 131 a and 131 b. The embodiment of this drawing is similar tothe embodiment of FIG. 16 except that a first order power calibrationunit 1316 a and a second order power calibration unit 1316 b are furtherincluded.

The first order power calibration unit 1316 a calibrates the power ofthe symbols mapped by the first order symbol mapper 1315 a according tothe size of the constellation and outputs the calibrated symbols. Thesecond order power calibration unit 1316 b calibrates the power of thesymbols mapped by the second order symbol mapper 1315 b according to thesize of the constellation and outputs the calibrated symbols.Accordingly, although the symbol mapping method is changed in one PLP oris changed among a plurality of PLPs, if the power of the symbol by thesymbol mapping method is adjusted according to the size of theconstellation, signal reception performance of a receiver can beimproved.

The symbol merger 1317 merges the symbols calibrated by the powercalibration units 1316 a and 1316 b and outputs one symbol stream.

FIG. 18 is a view showing another embodiment of the symbol mapper. Inthe embodiment of this Figure, the symbol mapper includes the secondencoder 125 and the second interleaver 127 included in the coding andmodulation unit. That is, if this embodiment is used, the coding andmodulation unit may include only the first encoder 121, the firstinterleaver 123 and the second encoder 125.

The embodiment of the symbol mapper includes a bit stream parser 1311, afirst order bit interleaver 1312 a, a second order bit interleaver 1312b, a first order demux 1313 a, a second order demux 1313 b, a firstorder symbol mapper 1315 a, a second order symbol mapper 1315 b and asymbol merger 1317.

When the second encoder 125 performs LDPC error correction coding, thelength of the error-correction-coded block (e.g., the length of 64800bits and the length of 16200 bits) may vary according to an LDPC mode.If the bits included in the error-correction-coded block are mapped tothe symbols, the error correction capabilities of the bits included in acell word configuring the symbol may vary according to the locations ofthe bits. For example, the cell word which is the symbol may bedetermined according to the code rate of the error correction coding andthe symbol mapping method (whether the symbol mapping method is thehigher order symbol mapping method or the lower order symbol mappingmethod). If the error-correction-code is the LDPC, the error correctioncapabilities of the bits vary according to the locations of the bits inthe error-correction-coded block. For example, the reliabilities of thebits coded according to the characteristics of the H-matrix used in theirregular LDPC error correction coding method may vary according to thelocations of the bits. Accordingly, the order of the bits configuringthe cell word mapped to the symbol is changed such that the errorcorrection capabilities of the bits which are weak against the errorcorrection in the error-correction-coded block are adjusted and therobustness against the error in the bit level can be adjusted.

First, the second encoder 125, for example, performs the errorcorrection coding with respect to the stream included in one PLP by theLDPC error correction coding method.

The bit stream parser 1311 receives the service stream according to thePLP and splits the received service stream.

The first order bit interleaver 1312 a interleaves the bits included ina first bit stream of the split service streams. Similarly, the secondorder bit interleaver 1312 b interleaves the bits included in a secondbit stream of the split service streams.

The first order bit interleaver 1312 a and the second order bitinterleaver 1312 b may correspond to the second interleaver 127 used asan inner interleaver. The interleaving method of the first order bitinterleaver 1312 a and the second order bit interleaver 1312 b will bedescribed later.

The first order demux 1313 a and the second order demux 1313 bdemultiplex the bits of the bit streams interleaved by the first orderbit interleaver 1312 a and the second order bit interleaver 1312 b. Thedemuxs 1313 a and 1313 b divide the input bit stream into sub bitstreams which will be mapped to a real axis and an imaginary axis of aconstellation and output the sub bit streams. The symbol mappers 1315 aand 1315 b map the sub bit streams demultiplexed by the demuxs 1313 aand 1313 b to the corresponding symbols.

The bit interleavers 1312 a and 1312 b and the demuxs 1313 a and 1313 bmay combine the characteristics of the LDPC codeword and thecharacteristics of the constellation reliability of the symbol mappingaccording to the constellation. The detailed embodiment of the firstorder demuxs 1313 a and 1313 b will be described later.

The first order symbol mapper 1315 a performs first order symbolmapping, for example, higher order symbol mapping, and the second ordersymbol mapper 1315 b performs second order symbol mapping, for example,lower order symbol mapping. The first order symbol mapper 1315 a mapsthe sub bit streams output from the first order demux 1313 to thesymbols and the second order symbol mapper 1315 b maps the sub bitstreams output from the second order demux 1313 b to the symbols.

The symbol merger 1317 merges the symbols mapped by the first ordersymbol mapper 1315 a and the second order symbol mapper 1315 b to onesymbol stream and outputs the symbol stream.

As described above, in the LDPC, the error correction capabilities ofthe bits may be changed according to the locations of the bits in theerror-correction-coded block. Accordingly, if the bit interleaver andthe demux are controlled according to the characteristics of the LDPCencoder 125 so as to change the order of the bits configuring the cellword, the error correction capability in the bit level can be maximized.

FIG. 19 is a view showing another embodiment of each of the symbolmappers 131 a and 131 b. The embodiment of this drawing is similar tothe embodiment of FIG. 18 except that a first order power calibrationunit 1316 a and a second order power calibration unit 1316 b are furtherincluded.

The first order power calibration unit 1316 a calibrates the power ofthe symbols mapped by the first order symbol mapper 1315 a according tothe size of the constellation and outputs the calibrated symbols. Thesecond order power calibration unit 1316 b calibrates the power of thesymbols mapped by the second order symbol mapper 1315 b according to thesize of the constellation and outputs the calibrated symbols.Accordingly, although the symbol mapping scheme is changed in one PLP oris changed among a plurality of PLPs, if the power of the symbol isadjusted according to the size of the constellation, signal receptionperformance can be improved.

The symbol merger 1317 merges the symbols calibrated by the powercalibration units 1316 a and 1316 b and outputs one symbol stream.

FIG. 20 is a view showing the concept of interleaving of bits by the bitinterleaves 1312 a and 1312 b of FIGS. 18 and 19.

For example, input bits are stored in and read from a matrix-formedmemory having a predetermined number of rows and columns. When the inputbits are stored, first, the bits are stored in a first column in rowdirection, and, if the first column is filled up, the bits are stored inanother column in row direction. When the stored bits are read, the bitsare read in column direction and, if all the bits stored in a first roware read, the bits in another row are read in column direction. In otherword, when the bits are stored, the bits are stored row-wise such thatthe columns are filled up serially. And when the stored bits are read,the stored bits are read column-wise from the first row to last rowserially. In this Figure, MSB means a most significant bit and LSB meansa least significant bit.

In order to map the LDPC-error-correction-coded bits to the symbols inthe same length of error correction block unit at various code rates,the bit interleavers 1312 a and 1312 b may change the number of rows andcolumns of the memory according to the types of the symbol mappers 1315a and 1315 b.

FIG. 21 is a view showing an example of the number of rows and columnsof memories of the bit interleavers 1312 a and 1312 b according to thetypes of symbol mappers 1315 a and 1315 b. If the LDPC mode is thenormal mode.

For example, if the symbol mapper 1315 a maps the bits to 256QAMsymbols, the first order interleaver 1312 a interleaves the bits by amemory having 8100 rows and columns. If the symbols are mapped by 64QAM.the first order interleaver 1312 a interleaves the bits by a memoryhaving 10500 rows and 6 columns. If the symbols are mapped by 16QAM, thefirst order interleaver 1312 a interleaves the bits by a memory having16200 rows and 4 columns.

For example, if the symbol mappers 1315 a and 1315 b map the bits toHyb128-QAM symbols. the first order interleaver 1312 a interleaves thebits using a memory having 4860 rows and 8 columns, and the second orderinterleaver 1312 b interleaves the bits using a memory having 4320 rowsand 6 columns.

Similarly, if the symbol mappers 1315 a and 1315 b map the symbols byHyb32-QAM, the first order interleaver 1312 a interleaves the bits usinga memory having 6480 rows and 6 columns, and the second orderinterleaver 1312 b interleaves the bits using a memory having 6480 rowsand 4 columns.

FIG. 22 is a view showing an example of the number of rows and columnsof the memories of the bit interleavers 1312 a and 1312 b according tothe types of the symbol mappers 1315 a and 1315 b, if the LDPC mode isthe short mode.

For example, if the symbol mapper 1315 a maps the bits to 256QAMsymbols, the first order interleaver 1312 a interleaves the bits by amemory having 2025 rows and 8 columns. If the symbol mappers 1315 a and1315 b map the symbols by Hyb128-QAM, the first order interleaver 1312 ainterleaves the bits using a memory having 1215 rows and 8 columns, andthe second order interleaver 1312 b interleaves the bits using a memoryhaving 1080 rows and 6 columns.

If the bit interleaving is performed with respect to theerror-correction-coded block, the locations of the bits in theerror-correction-coded block may be changed.

FIG. 23 is a diagram showing the concept of another embodiment ofinterleaving of a bit interleaver. In the embodiment shown in thisdrawing, when bits are written in a memory, the bits are written in acolumn direction. When the written bits are read, the bits of thecircularly shifted locations are read in a row direction. In each row,the bits written in each row is circularly shifted. If the bits arewritten or read by a circular shift method with respect to the row orthe column of the memory, this is called twisted bit interleaving. Thisembodiment relates to the twisted bit interleaving method using a methodof reading the bits after the bits are shifted by one column in rowdirection. Instead of shifting the written bits in the memory, the pointfor reading bits in the memory or the point for writing bits in thememory can be shifted.

In this embodiment, N denotes the length of the error correction codedblock and C denotes the length of the column. When the bits are written,the bits are written in a first column (represented by a shadow) inorder of 1, 2, 3, 4, and C and the bits are written in a second columnin order of C+1, C+2, C+3, . . . .

The written bits are twisted in the row direction one column by onecolumn.

If the written bits are read, the twisted bits are read in the rowdirection. For example, in this embodiment, the bits are read in a firstrow in order of 1, C+1, and the bits are read in a second row in orderof X1, 2, C+2, . . . . (X1 is a bit in the first column of the secondrow). The bits are read by row by row and the circularly shifted bitsare read. Of course, instead of shifting the written bits in the memory,the point for reading bits written in the memory can be shifted.

FIG. 24 is a view showing another embodiment of bit interleaving. Inthis embodiment, N denotes the length of the error correction codedblock and C denotes the length of the column. When the bits are written,the bits are written in a first column in order of 1, 2, 3, 4, . . . ,C−1, and C and the bits are written in a second column in order of C+1,C+2, C+3, . . . .

The written bits are double-twisted in the row direction two columns bytwo columns. If the written bits are read, the bits circularly shiftedby two columns are read in the column direction in every row. Thismethod may be called a double twisted bit interleaving method.

FIG. 25 is a view showing another embodiment of bit interleaving. Inthis embodiment, N denotes the length of the error correction codedblock and C denotes the length of the column. The bits are written in afirst column in order of 1, 2, 3, 4, . . . , C−1, and C and the bits arewritten in a second column in order of C+1, C+2, C+3, . . . .

When the written bits are read, in a first region of the rows, the bitsmay be read by the twisted bit interleaving method.

In a second region of the rows, the bits may be read by the doubletwisted interleaving method.

In a third region of the rows, the bits may be read by the twisted bitinterleaving method.

If the bits are interleaved by at least one of the twisted bitinterleaving method and the double twisted interleaving method, the bitsin the error correction coded block can be more randomly mixed.

FIG. 26 is a view showing the concept of multiplexing of input bits ofthe demurs 1313 a and 1313 b.

The bit interleavers 1312 a and 1312 b interleave the input bits x0, x1,and xn−1 and output the interleaved bits. The interleaving method isalready described above.

The demurs 1313 a and 1313 b demultiplex the interleaved bit streams.The demultiplexing method may vary according to the code rate of theerror correction coding method and the symbol mapping method of thesymbol mapper. If the symbol method of the symbol mapper is QPSK, theinput bits, for example, are interleaved to two sub streams and thesymbol mapper maps the two sub streams to the symbols so as tocorrespond to the real axis and the imaginary axis of the constellation.For example, a first bit y0 of the demultiplexed first sub streamcorresponds to the real axis and a first bit y1 of the demultiplexedsecond sub stream corresponds to the imaginary axis.

If the symbol method of the symbol mapper is 16QAM, the input bits, forexample, are demultiplexed to four sub frames. The symbol mapper selectsthe bits included in the four sub streams and maps the selected bits tothe symbols so as to correspond to the real axis and the imaginary axisof the constellation.

For example, the bits y0 and y2 of the demultiplexed first and third substreams correspond to the real axis and the bits y1 and y3 of thedemultiplexed second and fourth sub streams correspond to the imaginaryaxis.

Similarly, if the symbol method of the symbol mapper is 64QAM, the inputbits may be demultiplexed to six bit streams. The symbol mapper maps thesix sub streams to the symbols so as to correspond to the real axis andthe imaginary axis of the constellation. For example, the demultiplexedfirst, third and fifth sub stream bits y0, y2 and y4 correspond to thereal axis and the demultiplexed second, fourth and sixth sub stream bitsy1, y3 and y6 correspond to the imaginary axis.

Similarly, if the symbol method of the symbol mapper is 256QAM, theinput bits may be demultiplexed to eight bit streams. The symbol mappermaps the eight sub streams to the symbols so as to correspond to thereal axis and the imaginary axis of the constellation. For example,first, the demultiplexed first, third fifth and seventh sub stream bitsy0, y2, y4 and y6 correspond to the real axis and the demultiplexedsecond, fourth, sixth and eighth sub stream bits y1, y3, y6 and y7correspond to the imaginary axis.

If the symbol mapper maps the symbols, the sub streams demultiplexed bythe demux may be mapped to the bit streams of the real axis and theimaginary axis of the constellation.

The above-described bit interleaving method, demultiplexing method andsymbol mapping method are exemplary and various methods may be used asthe method of selecting the bits in the sub streams such that the substreams demultiplexed by the demux may correspond to the real axis andthe imaginary axis of the constellation.

The cell word mapped to the symbols may vary according to any one of theerror-corrected bit streams according to the code rate, the method ofinterleaving the bit streams, the demultiplexing method and the symbolmapping method. The MSB of the cell word is higher than the LSB of thecell word in the reliability of the error correction decoding. Althoughthe reliability of the bit of a specific location of theerror-correction-coded block is low, the reliability of the bit can beimproved by the symbol demapping process if the bit of the cell word isarranged on the MSB or close to the MSB.

Accordingly, although the reliability of the bit coded according to thecharacteristics of the H-matrix used in the irregular LDPC errorcorrection coding method is changed, the bit can be robustlytransmitted/received by the symbol mapping and demapping process and thesystem performance can be adjusted.

FIG. 27 is a view showing an embodiment of demultiplexing an inputstream by the demux.

If the symbol mapping method is QPSK, two bits are mapped to one symboland the two bits of one symbol unit are demultiplexed in order of thebit indexes (indexes 0 and 1 of b).

If the symbol mapping method is 16QAM, 4 bits are mapped to one symboland the four bits of one symbol unit are demultiplexed according to thecalculating result of the modulo-4 of bit indexes (indexes 0, 1, 2 and 3of b).

If the symbol mapping method is 64QAM, 6 bits are mapped to one symboland the six bits of one symbol unit are demultiplexed according to thecalculating result of the modulo-6 of bit indexes (indexes 0, 1, 2, 3, 4and 5 of b).

If the symbol mapping method is 256QAM, 8 bits are mapped to one symboland the eight bits of one symbol unit are demultiplexed according to thecalculating result of the modulo-8 of bit indexes (indexes 0, 1, 2, 3,4, 5, 6 and 7 of b).

The demultiplexing order of the sub streams is exemplary and may bemodified.

FIG. 28 is a view showing an example of a demultiplexing type accordingto a symbol mapping method. The symbol mapping method includes QPSK,16QAM, 64QAM and 256QAM, and the demultiplexing type includes a firsttype to a sixth type.

The first type is an example in which the input bits sequentiallycorrespond to even-numbered indexes (0, 2, 4, 8, . . . ) (or the realaxis of the constellation) and sequentially correspond to odd-numberedindexes (1, 3, 5, 7, . . . ) (or the imaginary axis of theconstellation). Hereinafter, the bit demultiplexing of the first typemay be represented by a demultiplexing identifier 10 (a binary number of1010; the location of 1 is, the location of the MSB corresponding, tothe real axis and the imaginary axis of the constellation).

The second type is an example in which the demultiplexing is performedin reverse order of the first, type, that is, the LSB of the input bitssequentially correspond to even-numbered indexes (6, 4, 2, 0) (or thereal axis of the constellation) and odd-numbered indexes (1, 3, 5, 7, .. . ) (or the imaginary axis of the constellation). Hereinafter, the bitdemultiplexing of the second type may be represented by a demultiplexingidentifier 5 (a binary number of 0101).

The third type is an example in which the input bits are arranged suchthat the bits of the both ends of the codeword become the MSB. The inputbits are rearranged so as to fill the code word from the both ends ofthe code word. Hereinafter, the bit demultiplexing of the third type maybe represented by a demultiplexing identifier 9 (a binary number of1001).

The fourth type is an example in which the input bits are arranged suchthat a middle bit of the code word becomes the MSB. A bit of the inputbits is first filled in the middle location of the code word and theremaining bits are then rearranged toward the both ends of the code wordin order of the input bits. Hereinafter, the bit demultiplexing of thefourth type may be represented by a demultiplexing identifier 6 (abinary number of 0110).

The fifth type is an example in which the bits are demultiplexed suchthat a last bit of the code word becomes the MSB and a first bit thereofbecomes the LSB, and the sixth type is an example in which the bits arerearranged such that the first bit of the code word becomes the MSB andthe last bit thereof becomes the LSB. Hereinafter, the bitdemultiplexing of the fifth type may be represented by a demultiplexingidentifier 3 (a binary number of 0011), and the bit demultiplexing ofthe sixth type may be represented by a demultiplexing identifier 12 (abinary number of 1100).

As described above, the demultiplexing type may vary according to thesymbol mapping method or the code rate of the error correction codingmethod. That is, a different demultiplexing type may be used if thesymbol mapping method or the code rate is changed.

FIG. 29 is a view showing an embodiment of demultiplexing an input bitstream according to a demultiplexing type. This embodiment may includebit interleavers 1312 a and 1312 b, demuxs 1313 a and 1313 b and mappers1315 a and 1315 b.

The bit interleavers 1312 a and 1312 b interleave theerror-correction-coded PLP service streams. For example, the bitinterleavers 1312 a and 1312 b may perform the bit interleaving in theerror correction coding units according to the error, correction codingmode. The bit interleaving method is already described above.

The demuxs 1313 a and 1313 b may include first type demuxs 1313 a 1 and1313 b 1, and nth type demuxs 1313 a 2 and 1313 b 2. Here, n is aninteger. The methods of demultiplexing the bits by the n types of demuxsfollow the types shown in FIG. 17. For example, the first type demuxsmay correspond to the first type bit demultiplexing (1100) and thesecond type demux (not shown) may correspond to the second type bitdemultiplexing (0011). The nth type demux 1313 b demultiplexes the inputbit stream according to the nth type bit multiplexing (e.g., thedemultiplexing identifier 1100) and outputs the demultiplexed bitstream. Selectors 1313 a 3 and 1313 b 3 receive a demux selection signalof the demultiplexing type suitable for the input bits and output thedemultiplexed bit stream according to any one of the first type to thenth type and the demux selection signal. The demux selection signal mayvary according to the code rate of the error correction coding and thesymbol mapping method of the constellation. Accordingly, thedemultiplexing type may be determined according to the code rate of theerror correction coding method or/and the symbol mapping method of theconstellation. The detailed example according to, the symbols mapped tothe constellation or/and the code rate of the error correction codingaccording to the demux selection signal will be described later.

The mappers 1315 a and 1315 b may map the demultiplexed sub bit streamsto the symbols according to the demux selection signal and output themapped symbols.

FIG. 30 is a view showing a demultiplexing type which is determinedaccording to a code rate of the error correction coding and the symbolmapping method.

In the 4QAM symbol mapping method, even when the code rate cr of theLDPC error correction coding method is any one of ¼, ⅓, ⅖, ½, ⅗, ⅔, ¾,⅘, ⅚, 8/9 and 9/10, the bit stream can be demultiplexed according to allthe demultiplexing types (denoted by all).

In the 16QAM symbol mapping method, if the code rate of the LDPC errorcorrection coding method is ¼, ⅓, ⅖ and ½, the symbols can be mappedwithout performing the bit interleaving and the bit demultiplexing(denoted by No-Int and No-Demux). If the code rate of the errorcorrection coding is ⅗, the bit can be demultiplexed according to anyone of the demultiplexing identifiers 9, 10 and 12. If the code rate ofthe error correction coding is ⅔, ¾, ⅘, ⅚, 8/9 and 9/10, the input bitstream can be demultiplexed according to the demultiplexing identifier6.

In the 64QAM symbol mapping method, if the code rate of the LDPC errorcorrection coding is ¼, ⅓, ⅖ and ½, the symbols can be mapped withoutperforming the bit interleaving and the bit demultiplexing. If the coderate is ⅗, the bits can be demultiplexed according to any one of thedemultiplexing identifiers 9 and 10. If the code rate is ⅔, ¾, ⅘, ⅚, 8/9and 9/10, the bits can be demultiplexed according to the demultiplexingidentifier 6.

In the 256QAM symbol mapping method, if the code rate of the LDPC errorcorrection coding is ¼, ⅓, ⅖ and ½, the symbols can be mapped withoutperforming the bit interleaving and the bit demultiplexing. If the coderate is ⅗, the bits can be demultiplexed according to the demultiplexingidentifier 9: If the code rate is ⅔, ¾; ⅘, ⅚, 8/9 and 9/10, the bits canbe demultiplexed according to the demultiplexing identifier 6.

As described above, the bit demultiplexing type may vary according tothe code rate used for the error correction coding and the symbolmapping method. Accordingly, the error correction capability of a bitlocated on a specific location of the error-correction-coded block maybe adjusted by mapping the demultiplexed sub streams to the symbols.Accordingly it is possible to optimize the robustness in the bit level.

FIG. 31 is a view showing an example of expressing the demultiplexingmethod by an equation. For example, if the symbol mapping method isQPSK, the input bits (xi, xN/2+i) correspond to the demultiplexed bitsy0 and y1. If the symbol mapping method is 16QAM, the input bits

$\left( {X_{\frac{2N}{4} + i},X_{\frac{3N}{4} + i},X,X_{\frac{N}{4} + i}} \right)$

correspond to the demultiplexed bits y0, y1, y2 and y3.

If the symbol mapping method is 64QAM, the input bits

$\left( {X_{\frac{4N}{6} + i},X_{\frac{5N}{6} + i},X_{\frac{2N}{6} + i},X_{\frac{3N}{6} + i},X,X_{\frac{N}{6} + i}} \right)$

correspond to the demultiplexed bits y0, y1, y2, y3, y4 and'y5. If thesymbol mapping method is 256QAM, the input bits

$\left( {X_{\frac{6N}{8} + i},X_{\frac{7N}{8} + i},X_{\frac{4N}{8} + i},X_{\frac{5N}{8} + i},X_{\frac{2N}{8} + i},X_{\frac{3N}{8} + i},X,X_{\frac{N}{8} + i}} \right)$

correspond to the demultiplexed bits y0, y1, y2, y3, y4, y5, y6 and y7.

Here, N denotes the number of bits mapped to the symbols with respect tothe input of the bit interleaver.

FIG. 32 is a view showing an example of mapping a symbol by a symbolmapper. For example, in the QPSK symbol mapping method, the symbols onthe constellation correspond to the value of the bit y0 of thedemultiplexed first sub stream and the value of the bit y1 of thedemultiplexed second sub stream.

In the 16QAM, the real axis of the symbols on the constellationcorresponds to the bits of the demultiplexed first and third sub streams(bits separated from the location of the MSB by 0 and 2) and theimaginary axis thereof corresponds to the bits of the demultiplexedsecond and fourth sub streams (bits separated from the location of theMSB by 1 and 3).

In the 64QAM, the real axis of the symbols on the constellationcorresponds to the bits of the demultiplexed first, third, and fifth substreams (bits separated from the location of the MSB by 0, 2 and 4) andthe imaginary axis thereof corresponds to the bits of the demultiplexedsecond, fourth and sixth sub streams (bits separated from the locationof the MSB by 1, 3 and 5).

Accordingly, the bits configuring the symbol may be mapped to the cellword in the demultiplexing order. If the bits configuring the cell wordare demultiplexed, the MSB and the LSB of the cell word are changed andthe robustness of the bits can be adjusted although the reliabilities ofthe LDPC error-correction-coded bits vary according to the locations.

FIG. 33 is a block diagram illustrating a MIMO/MISO encoder according toan embodiment of the present invention. The MIMO/MISO encoder encodesthe input data using the MIMO/MISO encoding scheme, and outputs theencoded data to several paths. If a signal reception end receives thesignal transmitted to the several paths from one or more paths, it isable to acquire a gain (also called a diversity gain, a payload gain, ora multiplexing gain).

The MIMO/MISO encoder 140 encodes service data of each path generatedfrom the frame builder 130, and outputs the encoded data to the A numberof paths corresponding to the number of output antennas.

FIG. 34 is a block diagram illustrating a modulator according to anembodiment of the present invention. The modulator includes a firstpower controller (PAPR Reduce1) 151, a time-domain transform unit (IFFT)153, a second power controller (PAPR Reduce2) 157, and a guard-intervalinserter 159.

The first power controller 151 reduces a PAPR (Peak-to-Average PowerRatio) of data transmitted to the R number of signal paths in thefrequency domain.

The time-domain transform (IF Fr) unit 153 converts the receivedfrequency-domain signals into time-domain signals. For example, thefrequency-domain signals may be converted into the time-domain signalsaccording to the IFFT algorithm. Therefore, the frequency-domain datamay be modulated according to the OFDM scheme.

The second power controller (PAPR Reduce2) 157 reduces a PAPR(Peak-to-Average Power Ratio) of channel data transmitted to the Rnumber of signal paths in the time domain. In this case, a tonereservation scheme, and an active constellation extension (ACE) schemefor extending symbol constellation can be used.

The guard-interval inserter 159 inserts the guard interval into theoutput OFDM symbol, and outputs the inserted result. As described above,the above-mentioned embodiment can be carried out in each signal of theR number of paths.

FIG. 35 is a block diagram illustrating an analog processor 160according to an embodiment of the present invention. The analogprocessor 160 includes a digital-to-analog converter (DAC) 161, anup-conversion unit 163, and an analog filter 165.

The DAC 161 converts the input data into an analog signal, and outputsthe analog signal. The up-conversion unit 163 converts a frequencydomain of the analog signal into an RF area. The analog filter 165filters the RF-area signal, and outputs the filtered RF signal.

FIG. 36 is a block diagram illustrating an apparatus for receiving asignal according to an embodiment of the present invention. The signalreception apparatus includes a first signal receiver 210 a, an n-thsignal receiver 210 n, a first demodulator 220 a, an n-th demodulator220 n, a MIMO/MISO decoder 230, a frame parser 240, and a decodingdemodulator 250, and an output processor 260.

In the case of a reception signal according to the TFS signal framestructure, several services are multiplexed to R channels, and are thentime-shifted, such that the time-shifted result is transmitted.

The receiver may include at least one signal receiver for receiving aservice transmitted over at least one RF channel. The TFS signal frametransmitted to the R (where R is a natural number) number of RF channelscan be transmitted to a multi-path via the A number of antennas. The Aantennas have been used for the R RF channels, such that a total numberof antennas is R×A.

The first signal receiver 210 a is able to receive service datatransmitted via at least one path from among overall service datatransmitted via several RF channels. For example, the first signalreceiver 210 a can receive the transmission signal processed by theMIMO/MISO scheme via several paths.

The first signal receiver 210 a and the n-th signal receiver 210 n canreceive several service data units transmitted over n number of RFchannels from among several RF channels, as a single PLP. Namely, thisembodiment shows the signal reception apparatus capable ofsimultaneously receiving data of the R number of RF channels. Therefore,if this embodiment receives a single RF channel, only the first receiver210 a is needed.

The first demodulator 220 a and the n-th demodulator 220 n demodulatesignals received in the first and n-th signal receivers 210 a and 210 naccording to the OFDM scheme, and output the demodulated signals.

The MIMO/MISO decoder 230 decodes service data received via severaltransmission paths according to the MIMO/MISO decoding scheme, andoutputs the decoded service data to a single transmission path. If thenumber R of services transmitted over several transmission paths arereceived, the MIMO/MISO decoder 230 can output single PLP service datacontained in each of R services corresponding to the number of Rchannels. If P number of services are transmitted via the R number of RFchannels, and signals of individual RF channels are received via the Anumber of antennas, the receiver decodes the P number of services usinga total of (R×A) reception antennas.

The frame parser 240 parses the TFS signal frame including severalservices, and outputs the parsed service data.

The decoding demodulator 250 performs the error correction decoding onthe service data contained in the parsed frame, demaps the decodedsymbol data into bit data, and outputs the demapping-processed result.

The output processor 260 decodes a stream including the demapped bitdata, and outputs the decoded stream.

In the above-mentioned description, each of the frame parser 240, andthe decoding demodulator 250, and the output processor 260 receivesseveral service data units as many as the number of PLPs, and performssignal processing on the received service data.

FIG. 37 is a block diagram illustrating a signal receiver according toan embodiment of the present invention. The signal receiver may includea tuner 211, a down-converter 213, and an analog-to-digital converter(ADC) 215.

The tuner 211 performs hopping of some RF channels capable oftransmitting user-selected services in all RF channels when the PLP isincluded in several RF channels, and outputs the hopping result. Thetuner 211 performs hopping of RF channels contained in the TFS signalframe according to input RF center frequencies, and at the same timetunes corresponding frequency signals, such that it outputs the tunedsignals. If a signal is transmitted to A number of multi-paths, thetuner 211 performs the tuning to a corresponding RF channel, andreceives reception signals via the A number of antennas.

The down converter 213 performs down conversion of the RF frequency ofthe signal tuned by the tuner 211, and outputs the down-conversionresult. The ADC 215 converts an analog signal into a digital signal.

FIG. 38 is a block diagram illustrating a demodulator according to anembodiment of the present invention. The demodulator includes a framedetector 221, a frame synchronization unit 222, a guard-interval remover223, a frequency-domain transform unit (FFT) 224, a channel estimator225, a channel equalizer 226, and a signaling-information extractor 227.

If the demodulator acquires service data transmitted to a single PLPstream, the following signal demodulation will be carried out. Adetailed description thereof will hereinafter be described.

The frame detector 221 identifies a delivery system of a receptionsignal. For example, the frame detector 221 determines whether thereception signal is a DVB-TS signal or not. And, the frame detector 221may also determine whether a reception signal is a TFS signal frame ornot. The frame synchronization unit 222 acquires time- andfrequency-domain synchronization of the TFS signal frame.

The guide interval controller 223 removes a guard interval locatedbetween OFDM symbols from the time domain. The frequency-domainconverter (FFT) 224 converts a reception signal into a frequency-domainsignal using the FFT algorithm, such that it acquires frequency-domainsymbol data.

The channel estimator 225 performs channel estimation of a receptionchannel using a pilot symbol contained in symbol data of the frequencydomain. The channel equalizer 226 performs channel equalization ofreception data using channel information estimated by the channelestimator 225.

The signaling information extractor 227 can extract the signalinginformation of a physical layer established in the first and secondpilot signals contained in channel-equalized reception data.

FIG. 39 is a block diagram illustrating a MIMO/MISO decoder according toan embodiment of the present invention. The signal receiver and thedemodulator are designed to process a signal received in a single path.If the signal receiver and the demodulator receive PLP service dataproviding a single service via several paths of several antennas, anddemodulate the PLP service data, the MIMO/MIMO decoder 230 outputs thesignal received in several paths as service data transmitted to a singlePLP. Therefore, the MIMO/MISO decoder 230 can acquire a diversity gainand a multiplexing gain from service data received in a correspondingPLP.

The MIMO/MISO decoder 230 receives a multi-path transmission signal fromseveral antennas, and is able to decode a signal using a MIMO schemecapable of recovering each reception signal in the form of a singlesignal. Otherwise, the MIMO/MISO decoder 230 is able to recover a signalusing a MIMO scheme which receives the multi-path transmission signalfrom a single antenna and recovers the received multi-path transmissionsignal.

Therefore, if the signal is transmitted via the R number of RF channels(where R is a natural number), the MIMO/MISO decoder 230 can decodesignals received via the A number of antennas of individual RF channels.If the A value is equal to “1”, the signals can be decoded by the MISOscheme. If the A value is higher than “1”, the signals can be decoded bythe MIMO scheme.

FIG. 40 is a block diagram illustrating a frame parser according to anembodiment of the present invention. The frame parser includes a firstfrequency de-interleaver 241 a, a r-th frequency de-interleaver 241 r, aframe parser 243, a first time de-interleaver 245 a, a p-th timede-interleaver 245 p, a first symbol demapper 247 a, and a p-th symboldemapper. The value of “r” can be decided by the number of RF channels,and the value of “p” can be decided by the number of streamstransmitting PLP service data generated from the frame parser 243.

Therefore, if p number of services are transmitted to p number of PLPstreams over R number of RF channels, the frame parser includes the rnumber of frequency deinterleavers, the p number of timede-interleavers, and the p number of symbol demappers.

In association with a first RF channel, the first frequency interleaver241 a performs de-interleaving of frequency-domain input data, andoutputs the de-interleaving result.

The frame parser 243 parses the TFS signal frame transmitted to severalRF channels using scheduling information of the TFS signal frame, andparses PLP service data contained in the slot of a specific RF channelincluding a desired service. The frame parser 243 parses the TFS signalframe to receive specific service data distributed to several RFchannels according to the TFS signal frame structure, and outputsfirst-path PLP service data.

The first time de-interleaver 245 a performs de-interleaving of theparsed first-path PLP service data in the time domain. The first symboldemapper 247 a determines service data mapped to the symbol to be bitdata, such that it can output a PLP stream associated with thefirst-path PLP service data.

Provided that symbol data is converted into bit data, and each symboldata includes symbols based on the hybrid symbol-mapping scheme, the pnumber of symbol demappers, each of which includes the first symboldemapper, can determine the symbol data to be bit data using differentsymbol-demapping schemes in individual intervals of the input symboldata.

FIG. 41 is a view showing an embodiment of each of symbol demappers 247a and 247 p. The symbol demappers receive the streams corresponding tothe PLPs from the time interleavers 245 a and 245 p respectivelycorresponding to the symbol demappers.

Each of the symbol demappers 247 a and 247 p may include an errorcorrection block splitter 2471, a symbol splitter 2473, a first orderdemapper 2475 a, a second order demapper 2475 b and a bit stream merger2478.

The error correction block splitter 2471 may split the PLP streamreceived from the corresponding one of the time interleavers 245 a and245 p in the error correction block units. The error correction blocksplitter 2471 may split the service stream in the normal mode LDPC blockunit. In this case, the service stream may be split in a state in whichfour blocks according to the short mode (the block having the length of16200 bits) are treated as the error correction block of one blockaccording to the normal mode (the block having the length of 64800bits).

The symbol splitter 2473 may split the symbol stream in the split errorcorrection block according to the symbol mapping method of the symbolstream.

For example, the first order demapper 2475 a converts the symbolsaccording to the higher order symbol mapping method into the bits. Thesecond order demapper 2475 b converts the symbols according to the lowerorder symbol mapping method into the bits.

The bit stream merger 2478 may receive the converted bits and output onebit stream.

FIG. 42 is a view showing another embodiment of each of the symboldemappers 247 a and 247 p. The embodiment of this drawing is similar tothe embodiment of FIG. 41 except that a first order power calibrationunit 2474 a and a second order power calibration unit 2474 h are furtherincluded.

The first order power calibration unit 2474 a receives the symbols splitby the symbol splitter 2473, calibrates the power of the receivedsymbols according to the symbol mapping schemes, and outputs thecalibrated symbols. The power of the received symbols may have the powercalibrated according to the size of the constellation based on thesymbol mapping methods. The first order power calibration unit 2474 aconverts the power calibrated in accordance with the into the originalsymbol power of the constellation. The first order demapper 2475 a maydemap the symbols, of which the power is calibrated by the first orderpower calibration unit, to the bits.

Similarly, the second order power calibration unit 2474 b receives thesymbols split by the symbol splitter 2473, modified the calibrated powerof the received symbols to the original power according to the size ofthe constellation, and outputs the modified symbols.

FIG. 43 is a view showing another embodiment of each of the symboldemappers 247 a and 247 p. Each of the symbol demappers 247 a and 247 pmay include a symbol splitter 2473, a first order demapper 2474 a, asecond order demapper 2474 b, a first order mux 2475 a, a second ordermux 2475 b, a first order bit deinterleaver 2476 a, a second order bitdeinterleaver 2476 b and a bit stream merger 2478. By this embodiment,the embodiment of the decoding and demodulation unit of FIG. 33 includesa first decoder 253, a first deinterleaver 255 and a second decoder 257.

The symbol splitter 2473 may split the symbol stream of the PLPaccording to the method corresponding to the symbol mapping method.

The first order demapper 2474 a and the second order demapper 2474 bconvert the split symbol streams into bits. For example, the first orderdemapper 2474 a performs the symbol demapping of the higher order QAMand the second order demapper 2474 b performs the symbol demapping ofthe lower order QAM. For example, the first order demapper 2474 a mayperform the symbol demapping of 256QAM and the second order demapper2474 b may perform the symbol demapping of 64QAM.

The first order mux 2475 a and the second order mux 2475 b multiplex thesymbol-mapped bits. The multiplexing methods may correspond to thedemultiplexing methods described with reference to FIGS. 15 to 18.Accordingly, the demultiplexed sub streams may be converted into one bitstream.

The first order bit deinterleaver 2476 a deinterleaves the bit streamsmultiplexed by the first order mux 2475 a. The second order bitdeinterleaver 2476 b deinterleaves the bits multiplexed by the firstorder mux 2475 a. The deinterleaving method corresponds to the bitinterleaving method. The bit interleaving method is shown in FIG. 12.

The bit stream merger 2478 may merge the bit streams deinterleaved bythe bit interleavers 2476 a and 2476 b to one bit stream.

The first decoder 253 of the decoding and demodulation unit may errorcorrection decode the output bit stream according to the normal mode orthe short mode and the code rate according to the modes.

FIG. 44 is a view showing another embodiment of each of the symboldemappers 247 a and 247 p. The embodiment of this drawing is similar tothe embodiment of FIG. 43 except that a first order power calibrationunit 2474 a and a second order power calibration unit 2474 b are furtherincluded. The first order power calibration unit 2474 a and the secondorder power calibration unit 2474 b modify the calibrated powers of thesymbols according to the symbol mapping methods and output the modifiedsymbols to the symbol demappers 2475 a and 2475 b.

FIG. 45 is a view showing an embodiment of multiplexing thedemultiplexed sub stream. In this embodiment, the demappers 2474 a and2474 b decide the cell words including the bits. The muxs 2475 a and2475 b multiplex the decided cell words according to the mux selectionsignal. The demultiplexed cell words are input to any one of first muxs2475 a 2 and 2475 b 2 to nth muxs 2475 a 3 and 2475 b 3.

The first muxs 2475 a 2 and 2475 b 2 to the nth muxs 2475 a 3 and 2475 b3 change the order of the bits in the cell words input according to themux selection signal. The mux selection signal may be changed accordingto the code rate of the error correction coding or the symbol mappingmethod. In order to generate one stream and the bit streams delivered tothe muxs, the order of selecting the sub stream may be changed accordingto the mux selection signal.

The first demuxs 2475 a 1 and 2475 b 1 output the symbol-demapped hitstreams to any one of the first muxs 2475 a 2 and 2475 b 2 to the nthmuxs 2475 a 3 and 2475 b 3 according to the mux selection signal. Thefirst sub muxs 2475 a 1 and 2475 b 1 may receive the sub streamsmultiplexed by the first muxs 2475 a 2 and 2475 b 2 to the n-th muxs2475 a 3 and 2475 b 3 and output one stream, according to the muxselection signal.

The cell words including the changed bits are input to the bitinterleavers 2476 a and 2476 b, and the bit deinterleavers 2476 a and2476 b deinterleave the input bits and output the deinterleaved bits.

FIG. 46 is a block diagram illustrating a decoding demodulator accordingto an embodiment of the present invention. The decoding demodulator mayinclude several function blocks corresponding to the coding andmodulation unit. In this embodiment, the decoding demodulator of FIG. 16may include a first de-interleaver 251, a first decoder 253, a secondde-interleaver 255, and a second decoder 257. The second de-interleaver255 can be selectively contained in the decoding demodulator.

The first de-interleaver 251 acts as an inner de-interleaver, and isable to perform deinterleaving of the p-th PLP stream generated from theframe parser.

The first decoder 253 acts as an inner decoder, can perform errorcorrection of the de-interleaved data, and can use an error correctiondecoding algorithm based on the LDPC scheme.

The second de-interleaver 255 acts as an outer interleaver, and canperform deinterleaving of the error-correction-decoded data.

The second decoder 257 acts as an outer decoder. Data de-interleaved bythe second de-interleaver 255 or error-corrected by the first decoder253 is error-corrected again, such that the second decoder 257 outputsthe re-error-corrected data. The second decoder 257 decodes data usingthe error correction decoding algorithm based on the BCH scheme, suchthat it outputs the decoded data.

The first de-interleaver 251 and the second de-interleaver 255 are ableto convert the burst error generated in data contained in the PLP streaminto a random error. The first decoder 253 and the second decoder 257can correct errors contained in data.

The decoding demodulator shows operation processes associated with asingle PLP stream. If the p number of streams exist, the p number ofdecoding demodulators are needed, or the decoding demodulator mayrepeatedly decode input data p times.

FIG. 47 is a block diagram illustrating an output processor according toan embodiment of the present invention. The output processor may includep number of baseband (BB) frame parsers (251 a, 261 p), a first servicemerger 263 a, a second service merger 263 b, a first demultiplexer 265a, and a second demultiplexer 265 b.

The BB frame parsers (261 a, 261 p) remove BB frame headers from thefirst to p-th PLP streams according to the received PLP paths, andoutput the removed result. This embodiment shows that service data istransmitted to at least two streams. A first stream is an MPEG-2 TSstream, and a second stream is a GS stream.

The first service merger 263 a calculates the sum of service datacontained in payload of at least one BB frame, such that it outputs thesum of service data as a single service stream. The first demultiplexer255 a may demultiplex the service stream, and output the demultiplexedresult.

In this way, the second service merger 263 b calculates the sum ofservice data contained in payload of at least one BB frame, such that itcan output another service stream. The second demultiplexer 255 b maydemultiplex the GS-format service stream, and output the demultiplexedservice stream.

FIG. 48 is a block diagram illustrating an apparatus for transmitting asignal according to an embodiment of another embodiment of an embodimentof by the present invention. The signal transmission apparatus includesa service composer 310, a frequency splitter 320, and a transmitter 400.The transmitter 400 encodes or modulates a signal including a servicestream to be transmitted to each RF band.

The service composer 310 receives several service streams, multiplexesseveral service streams to be transmitted to individual RF channels, andoutputs the multiplexed service streams. The service composer 310outputs scheduling information, such that it controls the transmitter400 using the scheduling information, when the transmitter 400 transmitsthe PLP via several RF channels. By this scheduling information, theservice composer 310 modulates several service frames to be transmittedto the several RF channels by the transmitter 400, and transmits themodulated service frames.

The frequency splitter 320 receives a service stream to be transmittedto each RF band, and splits each service stream into severalsub-streams, such that the individual RF frequency bands can beallocated to the sub-streams.

The transmitter 400 processes the service streams to be transmitted toindividual frequency bands, and outputs the processed resultant streams.For example, in association with a specific service stream to betransmitted to the first RF channel, the first mapper 410 maps the inputservice stream data into symbols. The first interleaver 420 interleavesthe mapped symbols to prevent the burst error.

The first symbol inserter 430 can insert a signal frame equipped with apilot signal (e.g., a scatter pilot signal or a continual pilot signal)into the modulated signal.

The first modulator 440 modulates the data interleaved by the signalmodulation scheme. For example, the first modulator 440 can modulatesignals using the OFDM scheme.

The first pilot symbol inserter 450 inserts the first pilot signal andthe second pilot signal in the signal frame, and is able to transmit theTFS signal frame.

Service stream data transmitted to the second RF channel is transmittedto the TFS signal frame via several blocks 415, 425, 435, 445, and 455of different paths shown in the transmitter of FIG. 18.

The number of signal processing paths transmitted from the transmitter400 may be equal to the number of RF channels contained in the TFSsignal frame.

The first mapper 410 and the second mapper may respectively include thedemultiplexers 1313 a and 1313 b, and allow the locations of the MSB andthe LSB to be changed in the symbol-mapped cell word.

FIG. 49 is a block diagram illustrating an apparatus for receiving asignal according to another embodiment of the present invention. Thesignal reception apparatus may include a reception unit 510, asynchronization unit 520, a mode detector 530, an equalizer 540, aparameter detector 550, a de-interleaver 560, a demapper 570, and aservice decoder 580.

The reception unit 500 is able to receive signals of a first RF channelselected by a user from among the signal frame. If the signal frameincludes several RF channels, the reception unit 500 performs hopping ofthe several RF channels, and at the same time can receive a signalincluding the selected service frame.

The synchronization unit 510 acquires synchronization of a receptionsignal, and outputs the synchronized reception signal. The demodulator520 is able to demodulate the synchronization-acquired signal. The modedetector 530 can acquire a FFT mode (e.g., 2k, 4k, 8k FFT operationlength) of the second pilot signal using the first pilot signal of thesignal frame.

The demodulator 520 demodulates the reception signal under the FFT modeof the second pilot signal. The equalizer 540 performs channelestimation of the reception signal, and outputs the channel-estimationresultant signal. The de-interleaver 560 deinterleaves thechannel-equalized reception signal. The demapper 570 demaps theinterleaved symbol using the symbol demapping scheme corresponding tothe transmission-signal symbol mapping scheme (e.g., QAM).

The parameter detector 550 acquires physical parameter information(e.g., Layer-1 (L1) information) contained in the second pilot signalfrom the output signal of the equalizer 540, and transmits the acquiredphysical parameter information to the reception unit 500 and thesynchronization unit 510. The reception unit 500 is able to change theRF channel to another channel using network information detected by theparameter detector 550.

The parameter detector 550 outputs service-associated information,service decider 580 decodes service data of the reception signalaccording to the service-associated information from the parameterdetector 550, and outputs the decoded service data.

The demapper 570 may include the muxs 2475 a and 2475 b and output thebit stream obtained by restoring the order of the bits of which thelocations of the MSB and the LSB are changed according to the code rateof the error correction coding and the symbol mapping method.

Hereinafter, a method for modulating a first pilot signal of a signalframe having at least one RF band and a method and apparatus forreceiving the modulated first pilot signal will be described.

The time-interleaved PLP symbols are transmitted via regions, which aretemporally divided in the signal frame. The time-interleaved PLP symbolsmay be transmitted via regions, which are divided in the frequencydomain, if a plurality of RF bands exists. Accordingly, if the PLP istransmitted or received, a diversity gain can be obtained. An errorcorrection mode and a symbol mapping method may be changed according toservices corresponding to transport streams or may be changed in theservice.

A first pilot signal and a second pilot Signal are arranged at the startlocation of the signal frame having such characteristics, as a preamblesignal.

As described above, the first pilot signal included in the signal framemay include an identifier for identifying the signal frame having theabove-described structure. The first pilot signal may includeinformation about the transmission structure indicating whether or notthe signal frame is transmitted via multiple paths and information aboutan FFT mode of a signal following the first pilot signal. The receivercan detect the signal frame from the first pilot signal and obtain theinformation about the integral carrier frequency offset estimation andinformation about the FFT mode of the data symbol.

FIG. 50 is a view showing an embodiment of the structure of a firstpilot signal. A portion denoted by A is a useful portion of the firstpilot signal. B denotes the same cyclic prefix as a first portion of theportion A in the time domain and C denotes the same cyclic suffix as asecond portion of the portion A in the time region. The first portionmay be duplicated from the second half of the portion A and the secondportion may be duplicated from the first half of the portion A.

B and C can be respectively obtained by duplicating the first portionand the second portion and frequency shifting the duplicated portions. Arelationship between B or C and A is as follows.

B=one part(A)·e^(j2πf) ^(SH) ^(t)

C=another part(A)·e^(j2πf) ^(SH) ^(t)  [Equation 1]

In the above equation, SH denotes a shift unit of the frequency shift.Accordingly, the frequency shift values of the portions B and C may beinversely proportional to the lengths of the portions B and C.

If the first pilot signal is configured by frequency shifting the cyclicprefix (B) and the cyclic suffix (C), the probability that the datasymbol is erroneously detected to the preamble is low and theprobability that the preamble is erroneously detected is reduced,although the data symbols configuring the PLP and the symbolsconfiguring the preamble are modulated in the same FFT mode.

If continuous wave (CW) interference is included like an analog TVsignal, the probability that the preamble is erroneously detected due toa noise DC component generated in a correlation process, is reduced. Inaddition, if the size of the FFT applied to the data symbols configuringthe PLP is larger than that of the FFT applied to the preamble, preambledetection performance can be improved even in a delay spread channelhaving a length equal to or greater than that of the useful symbolportion A of the preamble. Since both the cyclic prefix (B) and thecyclic suffix (C) are used in the preamble, the fractional carrierfrequency offset can be estimated by the correlation process.

FIG. 51 is a view showing an embodiment of detecting a preamble signalshown in FIG. 50 and estimating a timing offset and a frequency offset.This embodiment may be included in the frame detector 221 or the framesynchronization unit 222.

This embodiment may include a first delay unit 601, a complex conjugatecalculation unit 603, a first multiplier 605, a second multiplier 607, afirst filter 611, a second delay unit 615, a third multiplier 609, asecond filter 613, a fourth multiplier 617, a peak search unit 619, anda phase measurement unit 621.

The first delay unit 601 may delay a received signal. For example, thefirst delay unit 601 may delay the received signal by the length of theuseful symbol portion (A) of the first pilot signal.

The complex conjugate calculation unit 603 may calculate the complexconjugate of the delayed first pilot signal and output the calculatedsignal.

The first multiplier 605 may multiply the signal output from the complexconjugate calculation unit 603 by the received signal and output themultiplied signal.

Since the first pilot signal includes the portions B and C obtained byfrequency-shifting the useful portion A, the respective correlationvalues are obtained by shifting the received signals by the respectivefrequency shift amounts. In the first pilot signal, the portion B is aportion which is frequency-shifted up or frequency-shifted down from theportion A, and C is a portion which is frequency-shifted up orfrequency-shifted down from the portion A.

For example, if the output of the complex conjugate calculation unit 603is used, the output of the first multiplier 605 may include thecorrelation result of B (or the complex conjugate of B) and A (or thecomplex conjugate of A).

The second multiplier 607 may multiply the signal output from the firstmultiplier 605 by the frequency shift amount (denoted by ej^(fsH)t)applied to the portion B and output the multiplied signal.

The first filter 611 performs a moving average during a predeterminedperiod with respect to the signal output from the second multiplier 607.The moving average portion may be the length of the cyclic prefix (B) orthe length of the cyclic suffix (C). In this embodiment, the firstfilter 611 may calculate an average of the signal included in the lengthof the portion B. Then, in the result output from the first filter 611,the correlation value of the portions A and C included in the portion,of which the average is calculated, substantially becomes zero and thecorrelation result of the portions B and A remains. Since the signal ofthe portion B is multiplied by the frequency shift, value by the secondmultiplier 607, it is equal to the signal obtained by duplicating thesecond half of the portion A.

The third multiplier 609 may multiply the signal output from the firstmultiplier 605 by the frequency shift amount (denoted by −ejfSHt)applied to the portion C and output the multiplied signal.

The second filter 613 performs a moving average during a predeterminedperiod with respect to the signal output from the third multiplier 609.The moving average portion may be the length of the cyclic prefix (B) orthe length of the cyclic suffix (C). In this embodiment, the secondfilter 613 may calculate the average of the signal included in thelength of the portion C. Then, in the result output from the secondfilter 613, the correlation value of the portions A and B included inthe portion, of which the average is calculated, substantially becomeszero and the correlation result of the portions C and A remains. Sincethe signal of the portion C is multiplied by the frequency shift valueby the third multiplier 609, it is equal to the signal obtained byduplicating the first half of the portion A.

The length TB of the portion of which the moving average is performed bythe first filter 611 and the second filter 613 is expressed as follows.

T _(B) =k/f _(SH)  [Equation 2]

where, k denotes an integer. In other words, the unit fSH of thefrequency shift used in the portions B and C may be decided by k/TB.

The second delay unit 615 may delay the signal output from the firstfilter 611. For example, the second delay unit 615 delays the signalfiltered by the first filter 611 by the length of the portion B andoutputs the delayed signal.

The fourth multiplier 617 multiplies the signal delayed by the seconddelay unit 615 by the signal filtered by the second filter 613 andoutputs the multiplied signal.

The peak search unit 619 searches for the location where a peak value isgenerated from the multiplied signal output from the fourth multiplier617 and outputs the searched location to the phase measurement unit 621.The peak value and the location may be used for the timing offsetestimation.

The phase measurement unit 621 may measure the changed phase using thepeak value and the location output from the peak search unit 619 andoutput the measured phase. The phase value may be used for thefractional carrier frequency offset estimation.

Meanwhile, an oscillator for generating the frequency used forperforming the frequency shift by the second multiplier 607 and thethird multiplier 609 may generate any phase error.

Even in this case, the fourth multiplier 617 can eliminate the phaseerror of the oscillator. The results output from the first filter 611and the second filter 613 and the result output from the fourthmultiplier 617 may be expressed by the following equation.

y _(MAF1) =∥a ₁(n)∥² ·e ^(j2πΔf+θ)

y _(MAF2) =∥a ₂(n)∥² ·e ^(j2πΔf+θ)

y _(prod) =∥a ₁(n)∥² ∥a ₂(n)∥² ·e ^(j2πΔf+θ)  [Equation 3]

where, y_(MAF1) and y_(MAF2) respectively denote the outputs of thefirst filter 611 and the second filter 613, and yProd denotes the outputof the fourth multiplier 617. In addition, a1 and a2 respectively denotethe levels of the correlation results and f and respectively denote thefrequency offset and the phase error of the oscillator.

Accordingly, y_(MAF1) and y_(MAF2) may include the phase errors of theoscillator having different signs, but the phase error of the oscillatoris eliminated in the result of the fourth multiplier 617. Accordingly,the frequency offset f can be estimated regardless of the phase error ofthe oscillator of the signal receiving apparatus.

The estimated frequency offset may be expressed by the followingequation.

f _(B) =∠y _(prod)/4π  [Equation 4]

where, the estimated frequency offset f is 0<=f<0.5.

FIG. 52 is a view showing another embodiment of the structure of thefirst pilot signal. In the first pilot signal, the frequency shift ofthe first half of the useful portion A is the cyclic prefix (B) and thefrequency shift of the second shift of the useful portion A is thecyclic suffix (C). The lengths of the useful portion A for generatingthe portions B and C may be, for example, ½ of the length of the portionA, and the lengths of B and C may be different.

FIG. 53 is a view showing an embodiment of detecting the first pilotsignal shown in FIG. 52 and measuring a timing offset and a frequencyoffset using the detected result. In this embodiment, for convenience ofdescription. B and C respectively denote the cyclic prefix and thecyclic suffix obtained by frequency-shifting ½ of the length of theportion A.

This embodiment includes a first delay unit 601, a complex conjugatecalculation unit 603, a first multiplier 605, a second multiplier 607, afirst filter 611, a second delay unit 615, a third multiplier 609, asecond filter 613, a fourth multiplier 617, a peak search unit 619, anda phase measurement unit 621. That is, this embodiment is equal to theembodiment of FIG. 51, but the features of the components may be changedaccording to the length of the portion A by which the portions B and Care generated. B denotes a portion frequency-shifted down from theportion A, and C denotes a portion frequency-shifted up from the portionA.

The first delay unit 601 may delay a received signal. For example, thefirst delay unit 601 may delay the received signal by ½ of the length ofthe useful symbol portion A of the first pilot signal.

The complex conjugate calculation unit 603 may calculate the complexconjugate of the delayed first pilot signal and output the calculatedsignal.

The first multiplier 605 may multiply the signal output from the complexconjugate calculation unit 603 by the received signal and output themultiplied signal.

The second multiplier 607 may multiply the signal output from the firstmultiplier 605 by the frequency shift amount (denoted by ejfSHt) appliedto the portion B and output the multiplied signal.

The first filter 611 performs a moving average during a predeterminedperiod with respect to the signal output from the second multiplier 607.The moving average portion may be the length of the cyclic prefix (B).In this embodiment, the first filter 611 may calculate the average ofthe signal included in the length of the portion B. Then, in the resultoutput from the first filter 611, the correlation value of the portionsA and C included in the portion, of which the average is calculated,substantially becomes zero and the correlation result of the portions Band A remains. Since the signal of the portion B is multiplied by thefrequency shift value by the second multiplier 607, it is equal to thesignal obtained by duplicating the second half of the portion A.

The third multiplier 609 may multiply the signal output from the firstmultiplier 605 by the frequency shift amount (denoted by −ejfSHt)applied to the portion C and output the multiplied signal.

The second filter 613 performs a moving average during a predeterminedperiod with respect to the signal output from the third multiplier 609.The moving average portion may be the length of the cyclic suffix (C).In this embodiment, the second filter 613 may calculate the average ofthe signal included in the length of the portion C. Then, in the resultoutput from the second filter 613, the correlation value of A and Bincluded in the portion, of which the average is calculated,substantially becomes zero and the correlation result of the portions Cand A remains. Since the signal of the portion C is multiplied by thefrequency shift value by the third multiplier 609, it is equal to thesignal obtained by duplicating the first half of the portion A.

The second delay unit 615 may delay the signal output from the firstfilter 611. For example, the second delay unit 615 delays the signalfiltered by the first filter 611 by the length of the portion B+1/2A andoutputs the delayed signal.

The fourth multiplier 617 multiplies the signal delayed by the seconddelay unit 615 by the signal filtered by the second filter 613 andoutputs the multiplied signal.

The peak search unit 619 searches for the location where a peak value isgenerated from the multiplied signal output from the fourth multiplier617 and outputs the searched location to the phase measurement unit 621.The peak value and the location may be used for the timing offsetestimation.

The phase measurement unit 621 may measure the changed phase using thepeak value and the location output from the peak search unit 619 andoutput the measured phase. The phase value may be used for thefractional carrier frequency offset estimation.

As described above, an oscillator for generating the frequency used forperforming the frequency shift by the second multiplier 607 and thethird multiplier 609 may generate any phase error. However, even in thisembodiment, the fourth multiplier 617 can eliminate the phase error ofthe oscillator.

The results output from the first filter 611 and the second filter 613and the result output from the fourth multiplier 617 may be expressed bythe following equation,

y _(MAF1) =∥a ₁(n)∥² ·e ^(j2πΔf+θ)

y _(MAF2) =∥a ₂(n)∥² ·e ^(j2πΔf+θ)

y _(prod) =∥a ₁(n)∥² ∥a ₂(n)∥² ·e ^(j2πΔf+θ)  [Equation 3]

where, y_(MAF1), and y_(MAF2) respectively denote the outputs of thefirst filter 611 and the second filter 613, and y_(Prod) denotes theoutput of the fourth multiplier 617. In addition, a1 and a2 respectivelydenote the levels of the correlation results and f and respectivelydenote the frequency offset and the phase error of the oscillator.

Accordingly, yMAF1 and yMAF2 may include the phase errors of theoscillator having different signs, but the phase error of the oscillatoris eliminated in the result of the fourth multiplier 617. Accordingly,the frequency offset f can be estimated regardless of the phase error ofthe oscillator of the signal receiving apparatus.

The estimated frequency offset may be expressed by the followingequation.

f _(B) =∠y _(prod)/2π

where, the estimated frequency offset f is 0<=f<1.

That is, phase aliasing may be generated in a range of 0.5<=f<1 in thefrequency offset estimated in [Equation 4], but phase aliasing is notgenerated in the frequency offset estimated in [Equation 6].Accordingly, the frequency offset can be more accurately measured. Thestructure of the first pilot signal may be used in the data symbol andthe second frequency signal. If such a structure is used, offsetestimation performance such as CW interference can be improved and thereception performance of the receiver can be improved.

FIG. 54 is a view showing an embodiment of detecting the first pilotsignal and measuring a timing offset and a frequency offset using thedetected result.

This embodiment includes a first delay unit 601, a third delay unit 602,a first complex conjugate calculation unit 603, a second complexconjugate calculation unit 604, a first multiplier 605, a fifthmultiplier 606, a second multiplier 607, a first filter 611, a seconddelay unit 615, a third multiplier 609, a second filter 613, a fourthmultiplier 617, a peak search unit 619, and a phase measurement unit621.

In this embodiment, the first delay unit 601 may delay a receivedsignal. For example, the first delay unit 601 may delay the receivedsignal by the length of the cyclic suffix.

The third delay unit 602 may delay the signal delayed by the first delayunit 601. For example, the third delay unit 602 further delays thesignal by a difference between the length of the cyclic prefix and thelength of the cyclic suffix.

The first complex conjugate calculation unit 603 may calculate thecomplex conjugate of the signal delayed by the third delay unit 602 andoutput the calculated signal. The second complex conjugate calculationunit 604 may calculate the complex conjugate of the signal delayed bythe first delay unit 601 and output the calculated signal.

The first multiplier 605 may multiply the signal output from the firstcomplex conjugate calculation unit 603 by the received signal and outputthe multiplied signal. The fifth multiplier 606 may multiply the complexconjugate calculated by the second complex conjugate calculation unit604 by the received signal and output the multiplied signal.

The second multiplier 607 may multiply the signal output from the firstmultiplier 605 by the frequency shift amount (denoted by e^(jfSHi))applied to the portion B and output the multiplied signal.

The first filter 611 performs a moving average during a predeterminedperiod with respect to the signal output from the second multiplier 607.The moving average portion may be the length of the useful portion (A)of the first pilot signal.

The third multiplier 609 may multiply the signal output from the secondmultiplier 604 by the frequency shift amount (denoted by −e^(jfSHt))applied to the portion C and output the multiplied signal.

The second filter 613 performs a moving average during a predeterminedperiod with respect to the signal output from the third multiplier 609.The moving average portion may be the length of the useful portion A ofthe first pilot signal.

The second delay unit 615 may delay the signal output from the firstfilter 611. For example, the second delay unit 615 delays the signalfiltered by the first filter 611 by the length of the useful portion (A)of the first pilot signal and outputs the delayed signal.

The fourth multiplier 617 multiplies the signal delayed by the seconddelay unit 615 by the signal filtered by the second filter 613 andoutputs the multiplied signal. The fourth multiplier 617 may eliminatethe phase error of the oscillator.

The operations of the peak search unit 619 and the phase measurementunit 621 are equal to those of the above-described embodiment. The peaksearch unit 619 searches for the location where a peak value isgenerated from the multiplied signal output from the fourth multiplier617 and outputs the searched location to the phase measurement unit 621.The peak value and the location may be used for the timing offsetestimation.

FIG. 55 is a view showing an embodiment of a method of transmitting asignal.

A service stream is converted to a PLP (S110). The PLP can be generatedby modulating a service stream such as a transport stream and a GSEpacket, in which error correction encoding and symbol mapping areperformed on the service stream. The modulated service stream can bedistributed in at least one signal frame and can be transmitted over atleast one physical channel as a PLP. For example, a process ofmodulating a service stream to a PLP can be performed by following stepsS110 a to S110 d.

A service stream such as a transport stream and a GSE packettransferring service is error-correction-coded (S110 a). An errorcorrection coding scheme may be changed according to the servicestreams.

An LDPC error correction coding scheme may be used as the errorcorrection coding scheme and the error correction coding may beperformed at various code rates. The bits which areerror-correction-coded according to a specific error correction coderate may be included in an error correction coded block according to theerror correction coding mode. If the error correction coding scheme isthe LDPC, a normal mode (64800 bits) and a short mode (16200 bits) maybe used.

The error-correction-coded service stream is interleaved (S110 b). Theinterleaving may be performed by differentiating the directions forwriting and reading the bits included in the error correction codedblock in and from a memory. The number of rows and the number of columnsof the memory may be changed according to the error correction codingmode. The interleaving may be performed in the unit of the errorcorrection coded blocks.

The interleaved bits of the service stream are mapped to symbols (S110c). A symbol mapping method may be changed according to service streamsor in the service stream. For example, as the symbol mapping method, ahigher order symbol mapping method and a lower order symbol mappingmethod may be used. When the symbols are mapped, the interleaved bits ofthe service stream may be demultiplexed according to the symbol mappingmethod or the code rate of the error correction code, and the symbolsmay be mapped using the bits included in the demultiplexed sub streams.Then, the sequence of the bits in the cell word mapped to the symbolsmay be changed.

The mapped symbols are interleaved (S110 d). The mapped symbols may beinterleaved in the unit of error correction coded blocks. Timeinterleavers 132 a and 132 b may interleave the symbols in the unit oferror correction coded blocks. That is, the service stream isinterleaved again in the symbol level.

The PLP converted as described above is allocated in at least one signalframe and a preamble including a first pilot signal is arranged in abeginning part of the signal frame (S150). The allocation of the PLP maybe described as follow.

The interleaved symbols of the service stream are split, the splitsymbols are allocated to a signal frame having at least one frequencyband and including slots which are temporally split in the frequencybands, and a preamble including a first pilot signal is arranged in astart portion of the signal frame. The interleaved symbols of theservice stream may configure the PLP with respect to the service streamfor providing the service. The symbols configuring the PLP may be splitand allocated to the signal frame. The PLP may be allocated to at leastone signal frame having at least one frequency band. If a plurality offrequency bands is arranged, the symbols configuring the PLP may bearranged in the slots shifted between the frequency bands. The bitsincluded in the service stream may be arranged in the signal frame inthe unit of interleaved error correction coded blocks.

The signal frame is converted into a time domain according to an OFDMscheme (S160).

The cyclic prefix obtained by frequency-shifting a first portion of anuseful portion of the first pilot signal and the cyclic suffix obtainedby frequency-shifting a second portion of the useful portion areinserted into the first pilot signal in the time domain (S170). If thepreamble is not inserted in the frequency domain, the preamble includingthe first pilot signal and the second pilot signal may be inserted inthe time domain. The first pilot signal of the time domain may includethe useful portion, the cyclic prefix of the first portion of the usefulportion and the cyclic suffix of the second portion of the usefulportion. The first portion may be a backmost portion or the foremostportion of the useful portion. The second portion may be the foremostportion or the backmost portion of the useful portion.

The signal frame including the first frame signal is transmitted over atleast one RF channel (S180).

Since the useful portion of the first pilot signal includes thefrequency-shifted cyclic prefix and cyclic suffix, the signal frame canbe clearly identified as the structure of the first pilot signal. Thetiming offset or the frequency offset may be estimated and compensatedfor using the structure of the first pilot signal.

FIG. 56 is a view showing an embodiment of a method of receiving asignal.

A signal is received from a specific frequency band transferring signalframes (S210). The signal frame may be transmitted over at least onefrequency band. The signal may be received from a specific frequencyband

From the received signal, a first pilot signal including a cyclic prefixobtained by frequency-shifting a first portion of an useful portion anda cyclic suffix obtained by frequency-shifting a second portion of theuseful portion is identified, and the signal frame including the PLPs isidentified is demodulated by the OFDM scheme using the first pilotsignal (S220). The demodulating process using information set in thefirst pilot signal will be described in detail later.

The identified signal frame is parsed (S230). The signal frame mayinclude at least one frequency band. In the signal frame, a first PLPincluding the error correction coded blocks of the symbols, to which theservice stream is mapped, may be allocated to OFDM symbols together witha second PLP including the error correction coded blocks of anotherservice stream. If the signal frame includes a plurality of frequencybands, the error correction coded blocks of the PLP may be allocated tothe OFDM symbols which are temporally shifted in the plurality offrequency bands.

A service can be obtained from the PLP of the parsed signal frame(S240), in which this process is described in steps S240 a to S240 c.

The symbols, to which the service stream is mapped, are deinterleavedfrom the parsed signal frame (S240 a). The deinterleaving may beperformed in the symbol level which the service stream is mapped to. Forexample, the time, deinterleavers 245 a and 245 b may deinterleave theerror correction coded blocks including the symbols, to which theservice stream is mapped.

Then, the deinterleaved symbols are demapped so as to obtain the servicestream (S240 b). When the symbols are demapped, a plurality of substreams obtained by demapping the symbols may be output, the output substreams may be multiplexed, and the error-correction-coded servicestream may be output. The multiplexing scheme may be changed accordingto the symbol mapping method and the error correction code rate. Thesymbol demapping method may be changed in one service stream oraccording to service streams.

The service stream is deinterleaved and the deinterleaved service streamis error-correction-coded (240 c).

According to an apparatus for transmitting and receiving a signal and amethod for transmitting and receiving a signal of an embodiment of thepresent invention, it is possible to readily detect and restore atransmitted signal. In addition, it is possible to improve the signaltransmission/reception performance of the transmitting/receiving system.

FIG. 57 is a flowchart illustrating an embodiment of identifying a firstpilot signal and estimating an offset in a demodulating process.

The first pilot signal includes the cyclic prefix obtained byfrequency-shifting the first portion of the useful portion thereof andthe cyclic suffix obtained by frequency-shifting the second portion ofthe useful portion thereof. The timing offset and the frequency offsetmay be calculated using the first pilot signal as follows.

The received signal is delayed (S311). For example, the delay portionmay be the useful portion of the first pilot signal or ½ of the usefulportion. Alternatively, the delay portion may be the length of thecyclic prefix or the length of the cyclic suffix.

The complex conjugate of the delayed signal is calculated (S313).

The complex conjugate of the received signal and the delayed signal aremultiplied (S315). The delayed signal multiplied by the complexconjugate may be the signal having the above-described length. If thedelay signal is the length of the cyclic prefix or the cyclic suffix,the complex conjugate of the delayed signal may be calculated.

The signal multiplied by the complex conjugate is inversely shiftedaccording to the frequency shift of the cyclic prefix (S317). That is,the signal multiplied by the complex conjugate is shifted by the inverseshift amount of the frequency shift amount of the cyclic prefix signal.That is, a signal which is frequency shifted up is frequency shifteddown (or the signal which is frequency shifted down is frequency shiftedup).

Then, an average is calculated with respect to the signal which isinversely shifted according to the frequency shift of the cyclic prefix(S319). The portion of which the average is calculated may be the lengthof the cyclic prefix or the length of the useful portion A of the firstpilot signal depending on the embodiments. Since the average iscalculated with respect to the signal having the same length along withthe received signal, the moving average value may be output along withthe received signal.

The signal of which the average is calculated is delayed (S321). Thedelay portion may be the sum of the length of the cyclic prefix and thelength of ½ of the useful period, the length of the cyclic prefix. orthe length of the useful portion A of the first pilot signal, accordingto the embodiment.

The signal multiplied in the step S315 is inversely shifted according tothe frequency shift of the cyclic suffix (S323). The signal multipliedby the complex conjugate is shifted by the inverse shift amount of thefrequency shift amount of the cyclic suffix signal. That is, a signalwhich is frequency shifted up is frequency shifted down (or the signalwhich is frequency shifted down is frequency shifted up).

An average is calculated with respect to the signal which is inverselyshifted according to the frequency shift of the cyclic suffix (S325).The moving average is performed with respect to the signal correspondingto the length of the calculated cyclic suffix or the length of theuseful portion of the first pilot signal according to the embodiments.

The signal delayed in the step S321 and the signal of which the averageis calculated in the step S325 are multiplied (S327).

A peak location of the multiplied result is searched for (S329) and thephase of the signal is measured using the peak (S331). The searched peakmay be used for estimating the timing offset and the measured phase maybe used for estimating the frequency offset.

In this flowchart, the length of the cyclic suffix, the length of thecyclic prefix and the frequency inverse shift amount may be changed.

According to the apparatus for transmitting and receiving the signal andthe method for transmitting and receiving the signal of the invention,if the data symbol configuring the PLP and the symbols configuring thepreamble are modulated in the same FFT mode, the probability that thedata symbol is detected by the preamble is low and the probability thatthe preamble is erroneously detected is reduced. If continuous wave (CW)interference is included like the analog TV signal, the probability thatthe preamble is erroneously detected by a noise DC component generatedat the time of correlation is reduced.

According to the apparatus for transmitting and receiving the signal andthe method for transmitting and receiving the signal of the invention,if the size of the FFT applied to the data symbol configuring the PLP islarger than that of the FFT applied to the preamble, the preambledetecting performance may be improved even in a delay spread channelhaving a length equal to or greater than that of the useful symbolportion A of the preamble. Since both the cyclic prefix (B) and thecyclic suffix (C) are used in the preamble, the fractional carrierfrequency offset can be estimated.

The disclosed structure of the pilot signal may not be used for a signalframe including the PLP, and if the pilot signal is used for any signalframe, the described effect can be taken.

MODE FOR THE INVENTION

The embodiments of the invention are described in the best mode of theinvention.

INDUSTRIAL APPLICABILITY

A method of transmitting/receiving a signal and an apparatus fortransmitting/receiving a signal of the present invention can be used inbroadcast and communication fields.

1-15. (canceled)
 16. A method of transmitting a broadcast signal, themethod comprising: encoding physical layer pipe (PLP) data fordelivering a service; mapping the encoded PLP data to symbols; buildinga signal frame based on the mapped symbols; modulating the signal frameaccording to an orthogonal frequency division multiplexing (OFDM)scheme; inserting a pilot symbol into a beginning part of the modulatedsignal frame; and transmitting the broadcast signal including the signalframe and the inserted pilot symbol, wherein the pilot symbol comprisesan effective portion, a cyclic prefix obtained by frequency-shifting afirst portion of the effective portion, and a cyclic suffix obtained byfrequency-shifting a second portion of the effective portion.
 17. Themethod of claim 16, wherein: the first portion is a foremost part of theeffective portion; and the second portion is an endmost part of theeffective portion.
 18. An apparatus for transmitting a broadcast signal,the apparatus comprising; means for encoding physical layer pipe (PLP)data for delivering a service; means for mapping the encoded PLP data tosymbols; means for building a signal frame based on the mapped symbols;means for modulating the signal frame according to an orthogonalfrequency division multiplexing (OFDM) scheme; means for inserting apilot symbol into a beginning part of the modulated signal frame; andmeans for transmitting the broadcast signal including the signal frameand the inserted pilot symbol, wherein the pilot symbol comprises aneffective portion, a cyclic prefix obtained by frequency-shifting afirst portion of the effective portion, and a cyclic suffix obtained byfrequency-shifting a second portion of the effective portion.
 19. Theapparatus of claim 18, wherein: the first portion is a foremost part ofthe effective portion; and the second portion is an endmost part of theeffective portion.
 20. A method of receiving a broadcast signal, themethod comprising: receiving the broadcast signal including a signalframe, wherein the signal frame includes a pilot symbol in a beginningpart of the signal frame and physical layer pipe (PLP) data symbols fordelivering a service, and wherein the pilot symbol comprises aneffective portion, a cyclic prefix obtained by frequency-shifting afirst portion of the effective portion, and a cyclic suffix obtained byfrequency-shifting a second portion of the effective portion;demodulating the broadcast signal according to orthogonal frequencydivision multiplexing (OFDM); and decoding the demodulated broadcastsignal.
 21. The method of claim 20, wherein: the first portion is aforemost part of the effective portion; and the second portion is anendmost part of the effective portion.
 22. The method of claim 21,wherein demodulating the broadcast signal comprises: correlating theforemost part and the cyclic prefix to output first correlated values;delaying the first correlated values; correlating the endmost part andthe cyclic suffix to output second correlated values; multiplying thedelayed correlated values and the second correlated values; andestimating a frequency offset of the received broadcast signal.
 23. Anapparatus for receiving a broadcast signal, the apparatus comprising:means for receiving the broadcast signal including a signal frame,wherein the signal frame includes a pilot symbol in a beginning part ofthe signal frame and physical layer pipe (PLP) data symbols fordelivering a service, and wherein the pilot symbol comprises aneffective portion, a cyclic prefix obtained by frequency-shifting afirst portion of the effective portion, and a cyclic suffix obtained byfrequency-shifting a second portion of the effective portion; means fordemodulating the broadcast signal according to orthogonal frequencydivision multiplexing (OFDM); and means for decoding the demodulatedbroadcast signal.
 24. The apparatus of claim 23, wherein: the firstportion is a foremost part of the effective portion; and the secondportion is an endmost part of the effective portion.